Multi-path mitigation in rangefinding and tracking objects using reduced attenuation RF technology

ABSTRACT

A method and system for identification, tracking and locating in wireless communications and wireless networks. The method and system use reference and/or pilot signals that are present in wireless communications and wireless networks. The method and system might also use RTT, TOA and time-stamping measurements/techniques to determine one or more reference signals traveling time between the Base Station (eNB) or its functional equivalent and mobile device (UE) and or network device. Other wireless networks that do not use reference and/or pilot signals may employ one or more of the following types of alternate embodiments of present invention: 1) where a portion of frame is dedicated to the ranging signal/ranging signal elements; 2) where the ranging signal elements are embedded into (spread across) transmit/receive signals frame(s); and 3) where the ranging signal elements are embedded with the data. The method and system includes multi-path mitigations processor and multi-path mitigations techniques and algorithms which improve the track-locate accuracy. The method and system allow achieving increased accuracy by using multi-path mitigations processor and multi-path mitigations techniques and algorithms. The techniques of Digital Signal Processing and Software-Defined Radio are used.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of U.S. patent application Ser. No. 12/502,809, filed on Jul. 14, 2009, entitled METHODS AND SYSTEM FOR REDUCED ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, which is a continuation of U.S. patent application Ser. No. 11/610,595, filed on Dec. 14, 2006, entitled METHODS AND SYSTEM FOR REDUCED ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, which claims priority to U.S. provisional Patent Application No. 60/597,649 filed on Dec. 15, 2005, entitled METHOD AND SYSTEM FOR REDUCED ATTENUATION IN TRACKING OBJECTS USING MULTI-BAND RF TECHNOLOGY, which are incorporated by reference herein in their entirety.

U.S. patent application Ser. No. 12/502,809, filed on Jul. 14, 2009, entitled METHODS AND SYSTEM FOR REDUCED ATTENUATION IN TRACKING OBJECTS USING RF TECHNOLOGY, to which this application claims priority, is a non-provisional of U.S. provisional Patent Application No. 61/103,270, filed on 7 Oct. 2008, entitled METHODS AND SYSTEM FOR MULTI-PATH MITIGATION IN TRACKING OBJECTS USING REDUCED ATTENUATION RF TECHNOLOGY, which are incorporated by reference herein in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to wireless communications and wireless networks systems and systems for a Radio Frequency (RF)-based identification, tracking and locating of objects, including RTLS (Real Time Locating Service).

2. Background of the Related Art

RF-based identification and location-finding systems for determination of relative or geographic position of objects are generally used for tracking single objects or groups of objects, as well as for tracking individuals. Conventional location-finding systems have been used for position determination in an open outdoor environment. RF-based, Global Positioning System (GPS), and assisted GPSs are typically used. However, conventional location-finding systems suffer from certain inaccuracies when locating the objects in closed (i.e., indoor) environments, as well as outdoors. Although cellular wireless communication systems provide excellent data coverage in urban and most indoor environments, the position accuracy of these systems is limited by self-interference, multipath and non-line-of-sight propagation.

The indoor and outdoor location inaccuracies are due mainly to the physics of RF propagation, in particular, due to losses/attenuation of the RF signals, signal scattering and reflections. The losses/attenuation and scattering issues can be solved (see co-pending application Ser. No. 11/670,595) by employing narrow-band ranging signal(s) and operating at low RF frequencies, for example at VHF range or lower.

Although, at VHF and lower frequencies the multi-path phenomena (e.g., RF energy reflections), is less severe than at UHF and higher frequencies, the impact of the multi-path phenomena on location-finding accuracy makes location determination less reliable and precise than required by the industry. Accordingly, there is a need for a method and a system for mitigating the effects of the RF energy reflections (i.e., multi-path phenomena) in RF-based identification and location-finding systems that are employing narrow-band ranging signal(s).

As a rule, conventional RF-based identification and location-finding systems mitigating multipath by employing wide bandwidth ranging signals, e.g. exploiting wide-band signal nature for multi-path mitigation (see S. Salous, “Indoor and Outdoor UHF Measurements with a 90 MHz Bandwidth”, IEEE Colloquium on Propagation Characteristics and Related System Techniques for Beyond Line-of-Sight Radio, 1997, pp. 8/1-8/6). In addition, spatial diversity and/or antenna diversity techniques are used in some cases.

However, the spatial diversity may not be an option in many tracking-location applications because it leads to an increase in required infrastructure. Similarly, the antenna diversity has a limited value, because at lower operating frequencies, for example VHF, the physical size of antenna subsystem becomes too large. The case in point is the U.S. Pat. No. 6,788,199, where a system and method for locating objects, people, pets and personal articles is described.

The proposed system employs an antenna array to mitigate the multi-path. The optionally system operates at UHF in the 902-926 MHz band. It is well known that the linear dimension of the antenna is proportional to the wave length of an operating frequency. Also, the area of an antenna array is proportional to the square and volume to the cube of the linear dimensions ratio because in an antenna array the antennas are usually separated by ¼ or ½ wave length. Thus, at VHF and lower frequencies the size of the antenna array will significantly impact device portability.

On the other hand, because of a very limited frequency spectrum, the narrow bandwidth ranging signal does not lend itself into multi-path mitigation techniques that are currently used by conventional RF-based identification and location-finding systems. The reason is that the ranging signal distortion (i.e., change in the signal) that is induced by the multi-path is too small for reliable detection/processing in presence of noise. Also, because of limited bandwidth the narrow bandwidth receiver cannot differentiate between ranging signal Direct-Line-Of-Sight (DLOS) path and delayed ranging signal paths when these are separated by small delays, since the narrow bandwidth receiver lacks the required time resolution, which is proportional to the receiver's bandwidth (e.g., the narrow bandwidth has an integrating effect on the incoming signals).

Accordingly, there is a need in the art for a multi-path mitigation method and system for object identification and location-finding, which uses narrow bandwidth ranging signal(s) and operates in VHF or lower frequencies as well as UHF band frequencies and beyond.

The track and locate functionality need is primarily found in wireless networks. The multi-path mitigation methods and systems for object identification and location finding, described in co-pending application Ser. No. 12/502,809, can be utilized in most of the available wireless networks. However, certain wireless networks have communications standards/systems that require integration of the techniques into the wireless networks to fully benefit from various ranging and positioning signals that are described in co-pending application Ser. No. 12/502,809. Typically, these wireless systems can provide excellent data coverage over wide areas and most indoor environments. However, the position accuracy available with of these systems is limited by self-interference, multipath and non-line-of-sight propagation. As an example, the recent 3GPP Release 9 standardized positioning techniques for LTE (Long Term Evolution) standard has the following: 1) A-GNSS (Assisted Global Navigation Satellite System) or A-GPS (Assisted Global Positioning System) as the primary method; and 2) Enhanced Cell-ID and OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA), as fall-back methods. While these methods might satisfy the mandatory FCC E911 emergency location requirements, the accuracy, reliability and availability of these location methods fall short of the needs of LBS (Location Based Services) or RTLS system users, who require highly accurate locating within buildings, shopping malls, urban corridors, etc.

It is well known that the A-GNSS (A-GPS) accuracy is very good in open spaces but is very unreliable in urban/indoor environments. Also, Enhanced Cell-ID and OTDOA accuracy is impacted by the effects of multipath and other radio wave propagation phenomena.

There are other location methods, including ones that are based on received signal strength. However, RF propagation phenomenon make the signal strength vary 30 dB to 40 dB over the distance of a wavelength which, in wireless networks, can be significantly less than a meter. This severely impacts the accuracy and/or the reliability of methods based on received signal strength.

Accordingly, there is a need in the art for more accurate and reliable tracking and locating capability for wireless networks.

SUMMARY OF THE INVENTION

The present invention relates to a method and system for a Radio Frequency (RF)-based identification, tracking and locating of objects, including Real Time Locating Service (RTLS) that substantially obviates one or more of the disadvantages of the related art. The proposed (exemplary) method and system use a narrow bandwidth ranging locating signal(s). According to an exemplary embodiment, RF-based tracking and locating is implemented on VHF band, but could be also implemented on lower bands (HF, LF and VLF) as well as UHF band and higher frequencies. It employs multi-path mitigation method including techniques and algorithms. The proposed system can use software implemented digital signal processing and software defined radio technologies. Digital signal processing can be used as well.

The system of the exemplary embodiment can be constructed using standard FPGAs and standard signal processing hardware and software at a very small incremental cost to the device and overall system. At the same time the accuracy of the RF-based identification and location-finding systems that are employing narrow-band ranging signal/s can be significantly improved.

The transmitters and receivers for narrow bandwidth ranging/locating signal, for example VHF, are used to identify a location of a person or an object. Digital signal processing (DSP) and software defined radio (SDR) technologies can be used to generate, receive and process a narrow bandwidth ranging signal(s) as well as perform multi-path mitigation algorithms. The narrow bandwidth ranging signal is used to identify, locate and track a person or an object in a half-duplex, full duplex or simplex mode of operation. The Digital signal processing (DSP) and software defined radio (SDR) technologies are used in the multi-path mitigation processor to implement multi-path mitigation algorithms.

The approach described herein employs a multi-path mitigation processor and multi-path mitigation techniques/algorithms described in co-pending application Ser. No. 12/502,809 that increase the accuracy of tracking and locating system implemented by a wireless network. The present invention can be used in all wireless systems/networks and include simplex, half-duplex and full duplex modes of operation. The exemplary embodiment described below operates with wireless networks that employ OFDM modulation and/or its derivatives. Thus, the exemplary embodiment described below operates with LTE networks and it is also applicable to other wireless systems/networks.

The approach described herein is also applicable to other wireless networks, including WiMax, WiFi, and White Space. Other wireless networks that do not use reference and/or pilot signals may employ one or more of the following types of alternate modulation embodiments of present invention as described in co-pending application Ser. No. 12/502,809: 1) where a portion of frame is dedicated to the ranging signal/ranging signal elements as described in co-pending application Ser. No. 12/502,809; 2) where the ranging signal elements (see co-pending application Ser. No. 12/502,809) are embedded into transmit/receive signals frame(s); and 3) where the ranging signal elements (described in co-pending application Ser. No. 12/502,809) are embedded with the data.

These alternate embodiments employ multi-path mitigation processor and multi-path mitigation techniques/algorithms described in co-pending application Ser. No. 12/502,809 and can be used in all modes of operation: simplex, half-duplex and full duplex.

The integration of multi-path mitigation processor and multi-path mitigation techniques/algorithms described in co-pending application Ser. No. 12/502,809 with OFDM based wireless networks, and other wireless networks with reference/pilot signals, can be done with little or no incremental cost to the device and overall system. At the same time the location accuracy of the network and system will be significantly improved. As described in the exemplary embodiment, RF-based tracking and locating is implemented on 3GPP LTE cellular networks will significantly benefit from the localization of multi-path mitigation method/techniques and algorithms that are described in co-pending application Ser. No. 12/502,809 application. The proposed system can use software- or hardware-implemented digital signal processing.

Additional features and advantages of the invention will be set forth in the description that follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.

It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE ATTACHED FIGURES

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings:

FIG. 1 and FIG. 1A illustrate narrow bandwidth ranging signal frequency components, in accordance with the exemplary embodiment;

FIG. 2 illustrates exemplary wide bandwidth ranging signal frequency components.

FIG. 3A, FIG. 3B and FIG. 3C illustrate block diagrams of master and slave units of an RF mobile tracking and locating system, in accordance with the exemplary embodiment;

FIG. 4 illustrates an exemplary synthesized wideband base band ranging signal;

FIG. 5 illustrates elimination of signal precursor by cancellation, in accordance with the exemplary embodiment.

FIG. 6 illustrates precursor cancellation with fewer carriers, in accordance with the exemplary embodiment.

FIG. 7 illustrates a one-way transfer function phase.

FIG. 8 illustrates an exemplary embodiment location method.

FIG. 9 illustrates LTE reference signals mapping.

FIG. 10 illustrates an exemplary enhanced Cell ID+RTT locating technique.

FIG. 11 illustrates an exemplary OTDOA locating technique.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.

The present invention relates to a method and system for RF-based identification, tracking and locating of objects, including RTLS. According to the exemplary embodiment, the method and system employs a narrow bandwidth ranging signal. The exemplary embodiment operates in VHF band, but can be also used in HF, LF and VLF bands as well as UHF band and higher frequencies. It employs multi-path mitigation processor. Employing multi-path mitigation processor increases the accuracy of tracking and locating implemented by a system.

The exemplary embodiment includes small, highly portable base units that allow users to track, locate and monitor multiple persons and objects. Each unit has its own ID. Each unit broadcasts an RF signal with its ID, and each unit is able to send back a return signal, which can include its ID as well as voice, data and additional information. Each unit processes the returned signals from the other units and, depending on the triangulation or trilateration and/or other methods used, continuously determines their relative and/or actual locations. The preferred embodiment can also be easily integrated with products such as GPS devices, smart phones, two-way radios and PDAs. The resulting product will have all of the functions of the stand-alone devices while leveraging the existing display, sensors (such as altimeters, GPS, accelerometers and compasses) and processing capacity of its host. For example, a GPS device with the device technology describe herein will be able to provide the user's location on a map as well as to map the locations of the other members of the group.

The size of the preferred embodiment based on an FPGA implementation is between approximately 2×4×1 inches and 2×2×0.5 inches, or smaller, as integrated circuit technology improves. Depending on the frequency used, the antenna will be either integrated into the device or protrude through the device enclosure. An ASIC (Application Specific Integrated Circuit) based version of the device will be able to incorporate the functions of the FPGA and most of the other electronic components in the unit or Tag. The ASIC-based stand-alone version of the product will result in the device size of 1×0.5×0.5 inches or smaller. The antenna size will be determined by the frequency used and part of the antenna can be integrated into the enclosure. The ASIC based embodiment is designed to be integrated into products can consist of nothing more than a chipset. There should not be any substantial physical size difference between the Master or Tag units.

The devices can use standard system components (off-the-shelf components) operating at multiple frequency ranges (bands) for processing of multi-path mitigation algorithms. The software for digital signal processing and software-defined radio can be used. The signal processing software combined with minimal hardware, allows assembling the radios that have transmitted and received waveforms defined by the software.

Co-pending application Ser. No. 11/670,595 discloses a narrow-bandwidth ranging signal system, whereby the narrow-bandwidth ranging signal is designed to fit into a low-bandwidth channel, for example using voice channels that are only several kilohertz wide (though some of low-bandwidth channels may extend into a few tens of kilohertz). This is in contrast to conventional location-finding systems that use channels from hundreds of kilohertz to tens of megahertz wide.

The advantage of this narrow-bandwidth ranging signal system is as follows: 1) at lower operating frequencies/bands, conventional location-finding systems ranging signal bandwidth exceeds the carrier (operating) frequency value. Thus, such systems cannot be deployed at LF/VLF and other lower frequencies bands, including HF. Unlike conventional location-finding systems, the narrow-bandwidth ranging signal system described in co-pending application Ser. No. 11/670,595 can be successfully deployed on LF, VLF and other bands because its ranging signal bandwidth is far below the carrier frequency value; 2) at lower end of RF spectrum (some VLF, LF, HF and VHF bands), e.g., up to UHF band, conventional location-finding systems cannot be used because the FCC severely limits the allowable channel bandwidth (12-25 kHz), which makes it impossible to use conventional ranging signals. Unlike conventional location-finding systems, the narrow-bandwidth ranging signal system's ranging signal bandwidth is fully compliant with FCC regulations and other international spectrum regulatory bodies; and 3) it is well known (see MRI: the basics, by Ray H. Hashemi, William G. Bradley . . . —2003) that independently of operating frequency/band, a narrow-bandwidth signal has inherently higher SNR (Signal-to-Noise-Ratio) as compared to a wide-bandwidth signal. This increases the operating range of the narrow-bandwidth ranging signal location-finding system independently of the frequency/band it operates, including UHF band.

Thus, unlike conventional location-finding systems, the narrow-bandwidth ranging signal location-finding system can be deployed on lower end of the RF spectrum—for example VHF and lower frequencies bands, down to LF/VLF bands, where the multipath phenomena is less pronounced. At the same time, the narrow-bandwidth ranging location-finding system can be also deployed on UHF band and beyond, improving the ranging signal SNR and, as a result, increasing the location-finding system operating range.

To minimize multipath, e.g., RF energy reflections, it is desirable to operate on VLF/LF bands. However, at these frequencies the efficiency of a portable/mobile antenna is very small (about 0.1% or less because of small antenna length (size) relative to the RF wave length). In addition, at these low frequencies the noise level from natural and manmade sources is much higher than on higher frequencies/bands, for example VHF. Together, these two phenomena may limit the applicability of location-finding system, e.g. its operating range and/or mobility/portability. Therefore, for certain applications where operating range and/or mobility/portability are very important a higher RF frequencies/bands may be used, for example HF, VHF, UHF and UWB.

At VHF and UHF bands, the noise level from natural and manmade sources is significantly lower compared to VLF, LF and HF bands; and at VHF and HF frequencies the multi-path phenomena (e.g., RF energy reflections) is less severe than at UHF and higher frequencies. Also, at VHF, the antenna efficiency is significantly better, than on HF and lower frequencies, and at VHF the RF penetration capabilities are much better than at UHF. Thus, the VHF band provides a good compromise for mobile/portable applications. On the other hand in some special cases, for example GPS where VHF frequencies (or lower frequencies) cannot penetrate the ionosphere (or get deflected/refracted), the UHF can be a good choice. However, in any case (and all cases/applications) the narrow-bandwidth ranging signal system will have advantages over the conventional wide-bandwidth ranging signal location-finding systems.

The actual application(s) will determine the exact technical specifications (such as power, emissions, bandwidth and operating frequencies/band). Narrow bandwidth ranging allows the user to either receive licenses or receive exemption from licenses, or use unlicensed bands as set forth in the FCC because narrow band ranging allows for operation on many different bandwidths/frequencies, including the most stringent narrow bandwidths: 6.25 kHz, 11.25 kHz, 12.5 kHz, 25 kHz and 50 kHz set forth in the FCC and comply with the corresponding technical requirements for the appropriate sections. As a result, multiple FCC sections and exemptions within such sections will be applicable. The primary FCC Regulations that are applicable are: 47 CFR Part 90—Private Land Mobile Radio Services, 47 CFR Part 94 personal Radio Services, 47 CFR Part 15—Radio Frequency Devices. (By comparison, a wideband signal in this context is from several hundred KHz up to 10-20 MHz.)

Typically, for Part 90 and Part 94, VHF implementations allow the user to operate the device up to 100 mW under certain exemptions (Low Power Radio Service being an example). For certain applications the allowable transmitted power at VHF band is between 2 and 5 Watts. For 900 MHz (UHF band) it is 1 W. On 160 kHz-190 kHz frequencies (LF band) the allowable transmitted power is 1 Watt.

Narrow band ranging can comply with many if not all of the different spectrum allowances and allows for accurate ranging while still complying with the most stringent regulatory requirements. This holds true not just for the FCC, but for other international organizations that regulate the use of spectrum throughout the world, including Europe, Japan and Korea.

The following is a list of the common frequencies used, with typical power usage and the distance the tag can communicate with another reader in a real world environment (see Indoor Propagation and Wavelength Dan Dobkin, WJ Communications, V1.4 Jul. 10, 2002):

915 MHz 100 mW 150 feet 2.4 GHz 100 mW 100 feet 5.6 Ghz 100 mW  75 feet

The proposed system works at VHF frequencies and employs a proprietary method for sending and processing the RF signals. More specifically, it uses DSP techniques and software-defined radio (SDR) to overcome the limitations of the narrow bandwidth requirements at VHF frequencies.

Operating at lower (VHF) frequencies reduces scatter and provides much better wall penetration. The net result is a roughly ten-fold increase in range over commonly used frequencies. Compare, for example, the measured range of a prototype to that of the RFID technologies listed above:

216 MHz 100 mw 700 feet

Utilizing narrow band ranging techniques, the range of commonly used frequencies, with typical power usage and the distance the tag communication range will be able to communicate with another reader in a real world environment would increase significantly:

From: To: 915 MHz 100 mW 150 feet 500 feet 2.4 GHz 100 mW 100 feet 450 feet 5.6 Ghz 100 mW  75 feet 400 feet

Battery consumption is a function of design, transmitted power and the duty cycle of the device, e.g., the time interval between two consecutive distance (location) measurements. In many applications the duty cycle is large, 10× to 1000×. In applications with large duty cycle, for example 100×, an FPGA version that transmits 100 mW of power will have an up time of approximately three weeks. An ASIC based version is expected to increase the up time by 10×. Also, ASICs have inherently lower noise level. Thus, the ASIC-based version may also increase the operating range by about 40%.

Those skilled in the art will appreciate that the exemplary embodiment does not compromise the system long operating range while significantly increases the location-finding accuracy in RF challenging environments (such as, for example, buildings, urban corridors, etc.)

Typically, tracking and location systems employ Track-Locate-Navigate methods. These methods include Time-Of-Arrival (TOA), Differential-Time-Of-Arrival (DTOA) and combination of TOA and DTOA. Time-Of-Arrival (TOA) as the distance measurement technique is generally described in U.S. Pat. No. 5,525,967. A TOA/DTOA-based system measures the RF ranging signal Direct-Line-Of-Site (DLOS) time-of-flight, e.g., time-delay, which is then converted to a distance range.

In case of RF reflections (e.g., multi-path), multiple copies of the RF ranging signal with various delay times are superimposed onto the DLOS RF ranging signal. A track-locate system that uses a narrow bandwidth ranging signal cannot differentiate between the DLOS signal and reflected signals without multi-path mitigation. As a result, these reflected signals induce an error in the estimated ranging signal DLOS time-of-flight, which, in turn, impacts the range estimating accuracy.

The exemplary embodiment advantageously uses the multi-path mitigation processor to separate the DLOS signal and reflected signals. Thus, the exemplary embodiment significantly lowers the error in the estimated ranging signal DLOS time-of-flight. The proposed multi-path mitigation method can be used on all RF bands. It can also be used with wide bandwidth ranging signal location-finding systems. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH(Frequency Hopping).

Additionally, noise reduction methods can be applied in order to further improve the method's accuracy. These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc. The remnants of the multi-path interference error can be further reduced by applying the post-processing techniques, such as, maximum likelihood estimation (like.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc.

The exemplary embodiment can be used in systems with simplex, half-duplex and full duplex modes of operation. Full-duplex operation is very demanding in terms of complexity, cost and logistics on the RF transceiver, which limits the system operating range in portable/mobile device implementations. In half-duplex mode of operation the reader (often referred to as the “master”) and the tags (sometimes also referred to as “slaves” or “targets”) are controlled by a protocol that only allows the master or the slave to transmit at any given time.

The alternation of sending and receiving allows a single frequency to be used in distance measurement. Such an arrangement reduces the costs and complexity of the system in comparison with full duplex systems. The simplex mode of operation is conceptually simpler, but requires a more rigorous synchronization of events between master and target unit(s), including the start of the ranging signal sequence.

In present invention the narrow bandwidth ranging signal multi-path mitigation processor does not increase the ranging signal bandwidth. It uses different frequency components, advantageously, to allow propagation of a narrow bandwidth ranging signal. Further ranging signal processing can be carried out in the frequency domain by way of employing super resolution spectrum estimation algorithms (MUSIC, rootMUSIC, ESPRIT) and/or statistical algorithms like RELAX, or in time-domain by assembling a synthetic ranging signal with a relatively large bandwidth and applying a further processing to this signal. The different frequency component of narrow bandwidth ranging signal can be pseudo randomly selected, it can also be contiguous or spaced apart in frequency, and it can have uniform and/or non-uniform spacing in frequency.

The embodiment expands multipath mitigation technology. The signal model for the narrowband ranging is a complex exponential (as introduced elsewhere in this document) whose frequency is directly proportional to the delay defined by the range plus similar terms whose delay is defined by the time delay related to the multipath. The model is independent of the actual implementation of the signal structure, e.g., stepped frequency, Linear Frequency Modulation, etc.

The frequency separation between the direct path and multipath is nominally extremely small and normal frequency domain processing is not sufficient to estimate the direct path range. For example a stepped frequency ranging signal at a 100 KHz stepping rate over 5 MHz at a range of 30 meters (100.07 nanoseconds delay) results in a frequency of 0.062875 radians/sec. A multipath reflection with a path length of 35 meters would result in a frequency of 0.073355. The separation is 0.0104792. Frequency resolution of the 50 sample observable has a native frequency resolution of 0.12566 Hz. Consequently it is not possible to use conventional frequency estimation techniques for the separation of the direct path from the reflected path and accurately estimate the direct path range.

To overcome this limitation the invention uses a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis. The subspace decomposition technology relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace. The theory behind the subspace decomposition methodology is that the projection of the observable onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal.

The super resolution spectrum estimation algorithms and RELAX algorithm are capable of distinguishing closely placed frequencies (sinusoids) in spectrum in presence of noise. The frequencies do not have to be harmonically related and, unlike the Digital Fourier Transform (DFT), the signal model does not introduce any artificial periodicity. For a given bandwidth, these algorithms provide significantly higher resolution than Fourier Transform. Thus, the Direct Line Of Sight (DLOS) can be reliably distinguished from other multi-paths (MP) with high accuracy. Similarly, applying the thresholded method, which will be explained later, to the artificially produced synthetic wider bandwidth ranging signal makes it possible to reliably distinguish DLOS from other paths with high accuracy.

In accordance with the exemplary embodiment, the Digital signal processing (DSP), can be employed by the multi-path mitigation processor to reliably distinguish the DLOS from other MP paths. A variety of super-resolution algorithms/techniques exist in the spectral analysis (spectrum estimation) technology. Examples include subspace based methods: MUltiple SIgnal Characterization (MUSIC) algorithm or root-MUSIC algorithm, Estimation of Signal Parameters via Rotational Invariance Techniques (ESPRIT) algorithm, Pisarenko Harmonic Decomposition (PHD) algorithm, RELAX algorithm, etc.

In all of the abovementioned super-resolution algorithms the incoming (i.e., received) signal is modeled as a linear combination of complex exponentials and their complex amplitudes of frequencies. In case of a multi-path, the received signal will be as follows:

$\begin{matrix} {{{r(t)} = {\beta \times {\mathbb{e}}^{{\mathbb{i}2\pi}\; f \times t}{\sum\limits_{k = 0}^{k = {L - 1}}\;{\alpha_{k} \times {\mathbb{e}}^{{- {\mathbb{i}2\pi}}\; f \times \tau_{\kappa}}}}}},} & (1) \end{matrix}$

where β×e^(i2πf×t) is the transmitted signal, f is the operating frequency, L is the number of multi-path components, and α_(K)=|α_(K)|×e^(jθ) ^(K) and τ_(K) are the complex attenuation and propagation delay of the K-th path, respectively. The multi-path components are indexed so that the propagation delays are considered in ascending order. As a result, in this model τ₀ denotes the propagation delay of the DLOS path. Obviously, the τ₀ value is of the most interest, as it is the smallest value of all τ_(K). The phase θ_(K) is normally assumed random from one measurement cycle to another with a uniform probability density function U (0,2π). Thus, we assume that α_(K)=const (i.e., constant value)

Parameters α_(K) and π_(K) are random time-variant functions reflecting motions of people and equipment in and around buildings. However, since the rate of their variations is very slow as compared to the measurement time interval, these parameters can be treated as time-invariant random variables within a given measurement cycle.

All these parameters are frequency-dependent since they are related to radio signal characteristics, such as, transmission and reflection coefficients. However, in the exemplary embodiment, the operating frequency changes very little. Thus, the above-mentioned parameters can be assumed frequency-independent.

Equation (1) can be presented in frequency domain as:

$\begin{matrix} {{{A(f)} = {\sum\limits_{k = 0}^{k = {L - 1}}\;{\alpha_{k} \times {\mathbb{e}}^{{- {{\mathbb{i}}{({2\pi \times \tau_{\kappa}})}}}f}}}},} & (2) \end{matrix}$

where: A(f) is complex amplitude of the received signal, (2π×τ_(K)) are the artificial “frequencies” to be estimated by a super-resolution algorithm and the operating frequency f is the independent variable; α_(K) is the K-th path amplitude.

In the equation (2) the super-resolution estimation of (2π×τ_(K)) and subsequently τ_(K) values are based on continuous frequency. In practice, there is a finite number of measurements. Thus, the variable f will not be a continuous variable, but rather a discrete one. Accordingly, the complex amplitude A(f) can be calculated as follows:

$\begin{matrix} {{{\hat{A}\left( f_{n} \right)} = {\sum\limits_{k = 0}^{k = {L - 1}}\;{\alpha_{k} \times {\mathbb{e}}^{{- {{\mathbb{i}}{({2\pi \times \tau_{\kappa}})}}} \times f_{n}}}}},} & (3) \end{matrix}$

where Â(f_(n)) are discrete complex amplitude estimates (i.e., measurements) at discrete frequencies f_(n).

In equation (3) Â(f_(n)) can be interpreted as an amplitude and a phase of a sinusoidal signal of frequency f_(n) after it propagates through the multi-path channel. Note that all spectrum estimation based super-resolution algorithms require complex input data (i.e. complex amplitude).

In some cases, it is possible to convert real signal data, e.g. Re(Â(f_(n))), into a complex signal (e.g., analytical signal). For example, such a conversion can be accomplished by using Hilbert transformation or other methods. However, in case of short distances the value τ₀ is very small, which results in very low (2π×τ_(K)) “frequencies”.

These low “frequencies” create problems with Hilbert transform (or other methods) i\ implementations. In addition, if only amplitude values (e.g., Re(Â(f_(n)))) are to be used, then the number of frequencies to be estimated will include not only the (2π×τ_(K)) “frequencies”, but also theirs combinations. As a rule, increasing the number of unknown frequencies impacts the accuracy of the super-resolution algorithms. Thus, reliable and accurate separation of DLOS path from other multi-path (MP) paths requires complex amplitude estimation.

The following is a description of a method and the multi-path mitigation processor operation during the task of obtaining complex amplitude Â(f_(n)) in presence of multi-path. Note that, while the description is focused on the half-duplex mode of operation, it can be easily extended for the full-duplex mode. The simplex mode of operation is a subset of the half-duplex mode, but would require additional events synchronization.

In half-duplex mode of operation the reader (often referred to as the “master”) and the tags (also referred to as “slaves” or “targets”) are controlled by a protocol that only allows the master or the slave to transmit at any given time. In this mode of operation the tags (target devices) serve as Transponders. The tags receive the ranging signal from a reader (master device), store it in the memory and then, after certain time (delay), re-transmit the signal back to the master.

An example of ranging signal is shown in FIG. 1 and FIG. 1A. The exemplary ranging signal employs different frequency components that are contiguous. Other waveforms, including pseudo random, spaced in frequency and/or time or orthogonal, etc. can be also used for as long as the ranging signal bandwidth remains narrow. In FIG. 1 the time duration T_(f) for every frequency component is long enough to obtain the ranging signal narrow-bandwidth property.

Another variation of a ranging signal with different frequency components is shown on FIG. 2. It includes multiple frequencies (f₁, f₂, f₃, f₄, f_(n)) transmitted over long period of time to make individual frequencies narrow-band. Such signal is more efficient, but it occupies in a wide bandwidth and a wide bandwidth ranging signal impacts the SNR, which, in turn, reduces the operating range. Also, such wide bandwidth ranging signal will violate FCC requirements on the VHF band or lower frequencies bands. However, in certain applications this wide-bandwidth ranging signal allows an easier integration into existing signal and transmission protocols. Also, such a signal decreases the track-locate time.

These multiple-frequency (f₁, f₂, f₃, f₄, f_(n)) bursts may be also contiguous and/or pseudo random, spaced in frequency and/or time or orthogonal, etc.

The narrowband ranging mode will produce the accuracy in the form of instantaneous wide band ranging while increasing the range at which this accuracy can be realized, compared to wide band ranging. This performance is achieved because at a fixed transmit power, the SNR (in the appropriate signal bandwidths) at the receiver of the narrow band ranging signal is greater than the SNR at the receiver of a wideband ranging signal. The SNR gain is on the order of the ratio of the total bandwidth of the wideband ranging signal and the bandwidth of each channel of the narrow band ranging signal. This provides a good trade-off when very rapid ranging is not required, e.g., for stationary and slow-moving targets, such as a person walking or running.

Master devices and Tag devices are identical and can operate either in Master or Transponder mode. All devices include data/remote control communication channels. The devices can exchange the information and master device(s) can remotely control tag devices. In this example depicted in FIG. 1 during an operation of a master (i.e., reader) multi-path mitigation processor originates the ranging signal to tag(s) and, after a certain delay, the master/reader receives the repeated ranging signal from the tag(s).

Thereafter, master's multi-path mitigation processor compares the received ranging signal with the one that was originally sent from the master and determines the Â(f_(n)) estimates in form of an amplitude and a phase for every frequency component f_(n). Note that in the equation (3) Â(f_(n)) is defined for one-way ranging signal trip. In the exemplary embodiment the ranging signal makes a round-trip. In other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader. Thus, this round-trip signal complex amplitude, which is received back by the master, can be calculated as follows: (4) |Â _(RT)(f _(n))|=|Â(f _(n))|² and ∠Â _(RT)(f _(n))=2×(∠Â(f _(n)))  (4)

There are many techniques available for estimating the complex amplitude and phase values, including, for example, matching filtering |Â(f_(n))| and ∠Â(f_(n)). According to the exemplary embodiment, a complex amplitude determination is based on |Â(f_(n))| values derived from the master and/or tag receiver RSSI (Received Signal Strength Indicator) values. The phase values ∠Â_(RT)(f_(n)) are obtained by comparing the received by a reader/master returned base-band ranging signal phase and the original (i.e., sent by reader/master) base band ranging signal phase. In addition, because master and tag devices have independent clock systems a detailed explanation of devices operation is augmented by analysis of the clock accuracy impact on the phase estimation error. As the above description shows, the one-way amplitude |Â(f_(n))| values are directly obtainable from target/slave device. However, the one-way phase ∠Â(f_(n)) values cannot be measured directly.

In the exemplary embodiment, the ranging base band signal is the same as the one depicted in FIG. 1. However, for the sake of simplicity, it is assumed herein that the ranging base band signal consists of only two frequency components each containing multiple periods of cosine or sine waves of different frequency: F₁ and F₂. Note that F₁=f₁ and F₂=f₂. The number of periods in a first frequency component is L and the number of periods in a second frequency component is P. Note that L may or may not be equal to P, because for T_(f)=constant each frequency component can have different number of periods. Also, there is no time gap between each frequency component, and both F₁ and F₂ start from the initial phase equal to zero.

FIGS. 3A, 3B and 3C depict block diagrams of a master or a slave unit (tag) of an RF mobile tracking and locating system. F_(OSC) refers to the frequency of the device system clock (crystal oscillator 20 in FIG. 3A). All frequencies generated within the device are generated from this system clock crystal oscillator. The following definitions are used: M is a master device (unit); AM is a tag (target) device (unit). The tag device is operating in the transponder mode and is referred to as transponder (AM) unit.

In the preferred embodiment the device consists of the RF front-end and the RF back-end, base-band and the multi-path mitigation processor. The RF back-end, base-band and the multi-path mitigation processor are implemented in the FPGA 150 (see FIGS. 3B and 3C). The system clock generator 20 (see FIG. 3A) oscillates at: F_(OSC)=20 MHz; or ω_(OSC)=2π×20×10⁶. This is an ideal frequency because in actual devices the system clocks frequencies are not always equal to 20 MHz: F_(OSC) ^(M)=F_(OSC)γ^(M); F_(OSC) ^(AM)=F_(OSC)γ^(AM).

Note that

${\gamma^{M} = \frac{F_{OSC}^{M}}{F_{OSC}}},\mspace{14mu}{\gamma^{AM} = \frac{F_{OSC}^{AM}}{F_{OSC}}},\mspace{14mu}{{{and}\mspace{14mu}\beta^{M}} = \frac{1}{\gamma^{M}}},\mspace{14mu}{\beta^{AM} = \frac{1}{\gamma^{AM}}}$

It should be noted that other than 20 MHz F_(OSC) frequencies can be used without any impact on system performance.

Both units' (master and tag) electronic makeup is identical and the different modes of operations are software programmable. The base band ranging signal is generated in digital format by the master' FPGA 150, blocks 155-180 (see FIG. 2B). It consists of two frequency components each containing multiple periods of cosine or sine waves of different frequency. At the beginning, t=0, the FPGA 150 in a master device (FIG. 3B) outputs the digital base-band ranging signal to its up-converter 50 via I/Q DACs 120 and 125. The FPGA 150 starts with F₁ frequency and after time T₁ start generating F₂ frequency for time duration of T₂.

Since crystal oscillator's frequency might differ from 20 MHz the actual frequencies generated by the FPGA will be F₁γ^(M) and F₂γ^(M). Also, time T₁ will be T₁β^(M) and T₂ will be T₂β^(M). IT is also assumed that T₁, T₂, F₁, F₂ are such that F₁γ^(M)*T₁β^(M)=F₁T₁ and F₂γ^(M)*T₂β^(M)=F₂T₂, where both F₁T₁ & F₂T₂ are integer numbers. That means that the initial phases of F₁ and F₂ are equal to zero.

Since all frequencies are generated from the system crystal oscillator 20 clocks, the master' base-band I/Q DAC(s) 120 and 125 outputs are as follows: F₁=γ^(M) 20×10⁶×K_(F) ₁ and F₂=γ^(M) 20×10⁶ ×K_(F) ₂ , where K_(F) ₁ and K_(F) ₂ are constant coefficients. Similarly, the output frequencies TX_LO and RX_LO from frequency synthesizer 25 (LO signals for mixers 50 and 85) can be expressed through constant coefficients. These constant coefficients are the same for the master (M) and the transponder (AM)—the difference is in the system crystal oscillator 20 clock frequency of each device.

The master (M) and the transponder (AM) work in a half-duplex mode. Master's RF front-end up-converts the base-band ranging signal, generated by the multi-path mitigation processor, using quadrature up-converter (i.e., mixer) 50 and transmits this up-converted signal. After the base-band signal is transmitted the master switches from TX to RX mode using RF Front-end TX/RX Switch 15. The transponder receives and down-converts the received signal back using its RF Front-end mixer 85 (producing First IF) and ADC 140 (producing Second IF).

Thereafter, this second IF signal is digitally filtered in the Transponder RF back-end processor using digital filters 190 and further down-converted to the base-band ranging signal using the RF back-end quadrature mixer 200, digital I/Q filters 210 and 230, a digital quadrature oscillator 220 and a summer 270. This base-band ranging signal is stored in the transponder's memory 170 using Ram Data Bus Controller 195 and control logic 180.

Subsequently, the transponder switches from RX to TX mode using RF front-end switch 15 and after certain delay t_(RTX) begins re-transmitting the stored base-band signal. Note that the delay is measured in the AM (transponder) system clock. Thus, t_(RTX) ^(AM)=t_(RTX)β^(AM). The master receives the transponder transmission and down-converts the received signal back to the base-band signal using its RF back-end quadrature mixer 200, the digital I and Q filters 210 and 230, the digital quadrature oscillator 220 (see FIG. 3C).

Thereafter, the master calculates the phase difference between F₁ and F₂ in the received (i.e., recovered) base-band signal using multi-path mitigation processor arctan block 250 and phase compare block 255. The amplitude values are derived from the RF back-end RSSI block 240.

For improving the estimation accuracy it is always desirable to improve the SNR of the amplitude estimates from block 240 and phase difference estimates from block 255. In the preferred embodiment the multi-path mitigation processor calculates amplitude and phase difference estimates for many time instances over the ranging signal frequency component duration (T_(f)). These values, when averaged, improve SNR. The SNR improvement can be in an order that is proportional to √{square root over (N)}, where N is a number of instances when amplitude and phase difference values were taken (i.e., determined).

Another approach to the SNR improvement is to determine amplitude and phase difference values by applying matching filter techniques over a period of time. Yet, another approach would be to estimate the phase and the amplitude of the received (i.e., repeated) base band ranging signal frequency components by sampling them and integrating over period T≦T_(f) against the original (i.e., sent by the master/reader) base-band ranging signal frequency components in the I/Q form. The integration has the effect of averaging of multiple instances of the amplitude and the phase in the I/Q format. Thereafter, the phase and the amplitude values can be translated from the I/Q format to the |Â(f_(n))| and ∠Â(f_(n)) format.

Let's assume that at t=0 under master' multi-path processor control the master base-band processor (both in FPGA 150) start the base-band ranging sequence. φ_(FPGA) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (t)),t<T ₁β^(M) ,t<T ₁β^(M); φ_(FPGA) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(f) ₁ (T ₁β^(M))+K _(F) ₂ (t−T ₁β^(M))),t>T ₁β^(M), where T_(f)≧T₁β^(M). The phase at master's DAC(s) 120 and 125 outputs are as follows: φ_(DAC) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (t−t _(DAC) ^(M)))+φ_(DAC) ^(M)(0),t<T ₁β^(M) +t _(DAC) ^(M); φ_(DAC) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (T ₁β^(M))+K _(F) ₂ (t−T ₁β^(M) −t _(DAC) ^(M)))+φ_(DAC) ^(M)(0),t>T ₁β^(M) +t _(DAC) ^(M) Note that DACs 120 and 125 have internal propagation delay, t_(DAC) ^(M), that does not depend upon the system clock.

Similarly, the transmitter circuitry components 15, 30, 40 and 50 will introduce additional delay, t_(TX) ^(M), that does not depend upon the system clock.

As a result, the phase of the transmitted RF signal by the master can be calculated as follows: φ_(RF) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (t−t _(DAC) ^(M) −t _(TX) ^(M))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M)))+φ_(DAC) ^(M)(0)+φ_(SYN) _(—) _(TX) ^(M)(0), t<T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M); φ_(RF) ^(M)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (T ₁β^(M))+K _(F) ₂ (t−T ₁β^(M) −t _(DAC) −t _(TX) ^(M))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M)))+φ_(DAC) ^(M)(0)+φ_(SYN) _(—) _(TX) ^(M)(0), t>T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M)

The RF signal from the master (M) experiences a phase shift φ^(MULT) that is a function of the multi-path phenomena between the master and tag.

The φ^(MULT) values depend upon the transmitted frequencies, e.g. F₁ and F₂. The transponder (AM) receiver' is not able to resolve each path because of limited (i.e., narrow) bandwidth of the RF portion of the receiver. Thus, after a certain time, for example, 1 microsecond (equivalent to ˜300 meters of flight), when all reflected signals have arrived at the receiver antenna, the following formulas apply: φ_(ANT) ^(AM)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (t−t _(DAC) ^(M) −t _(TX) ^(M))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M)))+φ_(F) ₁ ^(MULT)+φ_(DAC) ^(M)(0)+φ_(SYN) _(—) _(TX) ^(M)(0), 10⁻⁶ <t<T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M); φ_(ANT) ^(AM)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (T ₁β^(M))+K _(F) ₂ (t−T ₁β^(M) −t _(DAC) ^(M) −t _(TX) ^(M))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M)))+φ_(F) ₂ ^(MULT)+φ_(DAC) ^(M)(0)+φ_(SYN) _(—) _(TX) ^(M)(0), t>T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M)+10⁻⁶

In the AM (transponder) receiver at the first down converter, element 85, an output, e.g. first IF, the phase of the signal is as follows: φ_(IF) _(—) ₁ ^(AM)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (t−t _(DAC) ^(M) −t _(TX) ^(M) −t _(RX) ^(AM))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M) −t _(RX) ^(AM)))−γ^(AM)×ω_(OSC)×(K _(SYN) _(—) _(RX) _(—) ₁ t)+φ_(F) ₁ ^(MULT)+φ_(SYN) _(—) ^(M)(0)−φ_(SYN) _(—) _(RX) _(—) ₁ ^(AM)(0),10⁻⁶ <t<T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M) +t _(TX) ^(AM); φ_(IF) _(—) ₁ ^(AM)(t)=γ^(M)×ω_(OSC)×(K _(F) ₁ (T ₁β^(M))+K _(f) ₂ (t−T ₁β^(M) −t _(DAC) ^(M) −t _(TX) ^(M) −t _(TX) ^(AM))+K _(SYN) _(—) _(TX)(t−t _(TX) ^(M) −t _(RX) ^(AM)))−γ^(AM)×ω_(OSC)×(K _(SYN) _(—) _(RX) _(—) ₁ t)+φ_(F2) ^(MULT)+φ_(SYN) _(—) _(TX) ^(M)(0)−φ_(SYN) _(—) _(RX) _(—) ₁ ^(AM)(0),t>T ₁β^(M) +t _(DAC) ^(M) +t _(TX) ^(M) +t _(RX) ^(AM)+10⁻⁶

Note that the propagation delay t in the receiver RF section (elements 15 and 60-85) does not depend upon the system clock. After passing through RF Front-end filters and amplifiers (elements 95-110 and 125) the first IF signal is sampled by the RF Back-end ADC 140. It is assumed that ADC 140 is under-sampling the input signal (e.g., first IF). Thus, the ADC also acts like a down-converter producing the second IF. The first IF filters, amplifiers and the ADC add propagation delay time. At the ADC output (second IF):

φ_(ADC)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM)) + K_(ADC)(t)) + φ_(F₁)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0), 10⁻⁶ < t < T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM); φ_(ADC)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM)) + K_(ADC)(t)) + φ_(F 2)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0), t > T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + 10⁻⁶

In the FPGA 150 the second IF signal (from the ADC output) is filtered by the RF Back-end digital filters 190 and further down-converted back to base-band ranging signal by the third down-converter (i.e., quadrature mixer 200, digital filters 230 and 210 and digital quadrature oscillator 220), summed in the summer 270 and is stored in the memory 170. At the third down-converter output (i.e., quadrature mixer):

φ_(BB)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) − t_(FIR)β^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM)) + K_(ADC)(t − t_(FIR)β^(AM)) + K_(SYN_RX_2)t) + φ_(F₁)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0), 10⁻⁶ < t < T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM); φ_(BB)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM)) + K_(ADC)(t − t_(FIR)β^(AM)) + K_(SYN_RX_2)t) + φ_(F₂)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0), t > T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + 10⁻⁶

Note that propagation delay t_(FIR) ^(AM)=t_(FIR)β^(AM) in the FIR section 190 does not depend upon the system clock.

After RX->TX delay the stored (in memory 170) base-band ranging signal from the master (M) is retransmitted. Note that RX->TX delay t_(RTX) ^(AM)=t_(RTX)β^(AM).

φ_(RF)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) − K_(SYN_TX)(t − t_(TX)^(AM))) + φ_(F₁)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM)(0), 10⁻⁶ < t < T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM); φ_(RF)^(AM)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) − K_(SYN_TX)(t − t_(TX)^(AM))) + φ_(F₂)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM)(0), t > T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM) + 10⁻⁶

By the time the signal from the transponder reaches the master' (M) receiver antenna MULT the RF signal from transponder (AM) experiences another phase shift φ^(MULT) that is a function of the multi-path. As discussed above, this phase shift happens after a certain time period when all reflected signals have arrived at the master' receiver antenna:

φ_(ANT)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) − K_(SYN_TX)(t − t_(TX)^(AM))) + 2 × φ_(F₁)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM)(0), 2 × 10⁻⁶ < t < T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM); φ_(ANT)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM)) − K_(SYN_TX)(t − t_(TX)^(AM))) + 2 × φ_(F₂)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM)(0), t > T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM) + 2 × 10⁻⁶

In the master receiver the signal from transponder goes through the same down-conversion process as in the transponder receiver. The result is the recovered base-band ranging signal that was originally sent by the master.

For the first frequency component F₁:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1  )  (t − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(t − t_(FIR)β^(M)) − K_(SYN_RX_2)(t)) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_TX)(t − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M))) + 2 × φ_(F₁)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM) − φ_(SYN_RX_1)^(M)(0) − φ_(ADC_CLK)^(M)(0) − φ_(SYN_RX_2)^(M)(0), 2 × 10⁻⁶ < t < T₁β^(M) + t_(DAC)^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM) + t_(RX)^(M) + t_(IF_1)^(M) + t_(ADC)^(M) + t_(FIR)β^(M);

For the second frequency component F₂:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − t_(DAC)^(M) − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_TX)(t − t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(t − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(t − t_(FIR)β^(M)) − K_(SYN_RX_2)(t)) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(ADC)(t − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_RX_2)(t − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_TX)(t − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M))) + 2 × φ_(F₂)^(MULT) + φ_(SYN_TX)^(M)(0) − φ_(SYN_RX_1)^(AM)(0) − φ_(ADC_CLK)^(AM)(0) − φ_(SYN_RX_2)^(AM)(0) + φ_(SYN_TX)^(AM) − φ_(SYN_RX_1)^(M)(0) − φ_(ADC_CLK)^(M)(0) − φ_(SYN_RX_2)^(M)(0), t > T₁β^(M) + t_(TX)^(M) + t_(RX)^(AM) + t_(IF_1)^(AM) + t_(ADC)^(AM) + t_(FIR)β^(AM) + t_(RTX)β^(AM) + t_(DAC)^(AM) + t_(TX)^(AM) + t_(RX)^(M) + t_(IF_1)^(M) + t_(ADC)^(M) + t_(FIR)β^(M) + 2 × 10⁻⁶

Substitutions: T _(D) _(—) _(M-AM) =t _(DAC) ^(M) +t _(DAC) ^(M) +t _(TX) ^(M) +t _(RX) ^(AM) +t _(IF) _(—) ₁ ^(AM) +t _(ADC) ^(AM) +t _(FIR)β^(AM) +t _(RTX)β^(AM) +t _(DAC) ^(AM) +t _(TX) ^(AM) +t _(RX) ^(M) +t _(IF) _(—) ₁ ^(M) +t _(ADC) ^(M) +t _(FIR)β^(M); where T_(D) _(—) _(M-AM) is the propagation delay through master (M) and transponder (AM) circuitry. φ_(BB) _(—) _(M-AM)(0)=φ_(SYN) _(—) _(TX) ^(M)(0)−φ_(SYN) _(—) _(RX) _(—) ₁ ^(AM)(0)−φ_(ADC) _(—) _(CLK) ^(AM)(0)−φ_(SYN) _(—) _(RX) _(—) ₂ ^(AM)(0)+φ_(SYN) _(—) _(TX) ^(AM)−φ_(SYN) _(—) _(RX) _(—) ₁ ^(M)(0)−φ_(ADC) _(—) _(CLK) ^(M)(0)−φ_(SYN) _(—) _(RX) _(—) ₂ ^(M)(0)=Const; where: φ_(BB) _(—) _(M-AM)(0) is the LO phase shift, at time t=0, from master (M) and transponder (AM) frequency mixers, including ADC(s). Also: K_(SYN) _(—) _(TX)=K_(SYN) _(—RX) _(—) ₁+K_(ADC)+K_(SYN) _(—) _(RX) _(—) ₂

First frequency component F₁:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − T_(D_M − AM)) − K_(SYN_TX)(t) + K_(SYN_RX_1)(t) − K_(ADC)(t) − K_(SYN_RX_2)(t) + K_(SYN_TX)(−t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(−t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(−t_(FIR)β^(M))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t) + K_(ADC)(t) + K_(SYN_RX_2)(t) − K_(SYN_TX)(t) + K_(SYN_RX_1)(t − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(ADC)(−t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(SYN_RX_2)(−t_(RTX)β^(AM) − t_(DAC)^(AM))K_(SYN_RX_1)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(ADC)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_RX_2)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_TX)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M))) + 2 × φ_(F₁)^(MULT) + φ_(BB_M − AM)(0), 2 × 10⁻⁶ < t < T₁β^(M) + T_(D_M − AM;)

First frequency component F₁ continued:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(t − T_(D_M − AM)) + K_(SYN_TX)(−t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(−t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(−t_(FIR)β^(M))γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(−t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(ADC)(−t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(SYN_RX_2)(−t_(RTX)β^(AM) − t_(DAC)^(AM))) + 2 × φ_(F₁)^(MULT) + φ_(BB_M − AM)(0), 2 × 10⁻⁶ < t < T₁β^(M) + T_(D_M − AM);

Second frequency component F₂:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − T_(D_M − AM)) − K_(SYN_TX)(t) + K_(SYN_RX_1)(t) − K_(ADC)(t) − K_(SYN_RX_2)(t) + K_(SYN_TX)(−t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(−t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(−t_(FIR)β^(M))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(t) + K_(ADC)(t) + K_(SYN_RX_2)(t) − K_(SYN_TX)(t) + K_(SYN_RX_1)(−t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM)) + K_(ADC)(−t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(SYN_RX_2)(−t_(RTX)β^(AM) − t_(DAC)^(AM))K_(SYN_RX_1)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(ADC)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) + K_(SYN_RX_2)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_TX)(−t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M))) + 2 × φ_(F₂)^(MULT) + φ_(BB_M − AM)(0), t > T₁β^(M) + T_(D_M − AM) + 2 × 10⁻⁶

Second frequency component F₂, continued:

φ_(BB_RECOV)^(M)(t) = γ^(M) × ω_(OSC) × (K_(F₁)(T₁β^(M)) + K_(F₂)(t − T₁β^(M) − T_(D_M − AM)) + K_(SYN_TX)(−t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(−t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(−t_(FIR)β^(M))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(−t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM)) + K_(ADC)(−t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(SYN_RX_2)(−t_(RTX)β^(AM) − t_(DAC)^(AM))) + 2 × φ_(F₂)^(MULT) + φ_(BB_M − AM)(0), t > T₁β^(M) + T_(D_M − AM) + 2 × 10⁻⁶

Further substituting:

α = γ^(M) × ω_(OSC) × (K_(SYN_TX)(−t_(TX)^(M) − t_(RX)^(AM) − t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM) − t_(TX)^(AM) − t_(RX)^(M) − t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(SYN_RX_1)(−t_(IF_1)^(M) − t_(ADC)^(M) − t_(FIR)β^(M)) − K_(ADC)(−t_(FIR)β^(M))) − γ^(AM) × ω_(OSC) × (K_(SYN_RX_1)(−t_(IF_1)^(AM) − t_(ADC)^(AM) − t_(FIR)β^(AM) − t_(RTX)β^(AM)) + K_(ADC)(−t_(FIR)β^(AM) − t_(RTX)β^(AM) − t_(DAC)^(AM)) + K_(SYN_RX_2)(−t_(RTX)β^(AM) − t_(DAC)^(AM))),

where α is a constant.

Then the final phase equations is:

$\begin{matrix} {{{{\varphi_{BB\_ RECOV}^{M}(t)} = {{\gamma^{M} \times \omega_{OSC} \times \left( {K_{F_{1}}\left( {t - T_{{D\_ M} - {AM}}} \right)} \right)} + {2 \times \varphi_{F_{1}}^{MULT}} + {\varphi_{{BB\_ M} - {AMN}}(0)} + \alpha}},{{{2 \times 10^{- 6}} < t < {{T_{1}\beta^{M}} + T_{{D\_ M} - {AM}}}};}}{{{\varphi_{BB\_ RECOV}^{M}(t)} = {{\gamma^{M} \times \omega_{OSC} \times \left( {{K_{F_{1}}\left( {T_{1}\beta^{M}} \right)} + {K_{F_{2}}\left( {t - {T_{1}\beta^{M}} - T_{{D\_ M} - {AM}}} \right)}} \right)} + {2 \times \varphi_{F_{2}}^{MULT}} + {\varphi_{{BB\_ M} - {AM}}(0)} + \alpha}},{t > {{T_{1}\beta^{M}} + T_{{D\_ M} - {AM}} + {2 \times 10^{- 6}}}}}} & (5) \end{matrix}$

From the equation (5):

${\angle{{\hat{A}}_{RT}\left( f_{n} \right)}} = \left\langle \begin{matrix} {{2 \times \varphi_{F_{1}}^{MULT}};{{2 \times \varphi_{F_{1}}^{MULT}} + {2 \times {\Delta\Phi}_{F_{1}/F_{2}}}};{{2 \times \varphi_{F_{1}}^{MULT}} +}} \\ {{2 \times {\Delta\Phi}_{F_{1}/F_{3}}};{{2 \times \varphi_{F_{1}}^{MULT}} + {2 \times {\Delta\Phi}_{F_{1}/F_{4}}}};\ldots\mspace{14mu};{{2 \times \varphi_{F_{1}}^{MULT}} + {2 \times {\Delta\Phi}_{F_{1}/F_{i}}}};} \end{matrix} \right\rangle$

where i=2, 3, 4 . . . ; and 2×ΔΦ_(F) ₁ _(/F) _(i) is equal to 2×(φ_(F) _(i) ^(MULT)−φ_(F) ₁ ^(MULT)).

For example, the difference 2×(φ_(F) ₂ ^(MULT)−φ_(F) ₁ ^(MULT)) at time instances t1 and t2:

2 × φ_(F₂)^(MULT) − 2 × φ_(F₁)^(MULT) = 2 × ΔΦ_(F₁/F₂) = φ_(BB_RECOV)^(M)(t₂) − φ_(BB_RECOV)^(M)(t₁) − γ^(M) × ω_(OSC) × [K_(F₁)(T₁β^(M)) + (K_(F₂)(t₂ − T₁β^(M) − T_(D_M − AM))) − (K_(F₁)(t₁ − T_(D_M − AM)))], 2 × 10⁻⁶ < t₁ < T₁β^(M) + T_(D_M − AM); t₂ > T₁β^(M) + T_(D_M − AM) + 2 × 10⁻⁶

To find 2×ΔΦ_(F) ₁ _(/F) ₂ difference we need to know T_(D) _(—) _(M-AM): T _(D) _(—) _(M-AM) =T _(LB) _(—) _(M)β^(M) +T _(LB) _(—) _(AM)β^(AM) +t _(RTX)β^(AM); T _(LB) _(—) _(M) =t _(DAC) ^(M) +t _(TX) ^(M) +t _(RX) ^(M) +t _(IF) _(—) ₁ ^(M) +t _(ADC) ^(M) +t _(FIR)β^(M) ;T _(LB) _(—) _(AM) =t _(DAC) ^(AM) +t _(TX) ^(AM) t _(RX) ^(AM) +t _(IF) _(—) ₁ ^(AM) +t _(ADC) ^(AM) +t _(FIR)β^(AM), where T_(LB) _(—) _(M) and T_(LB) _(—) _(AM) are propagation delays through the master (M) and transponder (AM) TX and RX circuitries that are measured by placing devices in the loop-back mode. Note that the master and the transponder devices can measure T_(LB) _(—) _(M) and T_(LB) _(—) _(AM) automatically; and we also know the t_(RTX) value.

From the above formulas and t_(RTX) value T_(D) _(—) _(M-AM) can be determined and consequently, for a given t₁, and t₂ the 2×ΔΦ_(F) ₁ _(/F) ₂ value can be found as follows:

$\begin{matrix} {{\left. {{{2 \times {\Delta\Phi}_{F_{1}/F_{2}}} = {{\varphi_{BB\_ RECOV}^{M}\left( t_{2} \right)} - {\varphi_{BB\_ RECOV}^{M}\left( t_{1} \right)} - {\gamma^{M} \times \omega_{OSC} \times \left\lbrack {{K_{F_{1}}\left( {T_{1}\beta^{M}} \right)} + {K_{F_{2}}t_{2}} - {K_{F_{2}}T_{1}\beta^{M}} - {K_{F\; 1}t_{1}} - {K_{F_{2}}T_{LB\_ M}\beta^{M}} + {K_{F_{1}}T_{LB\_ M}\beta^{M}} - {K_{F_{2}}\left( {{T_{LB\_ AM}\beta^{AM}\beta^{M}} + {t_{RTX}\beta^{M}}} \right)} + {K_{F_{1}}\left( {{T_{LB\_ AM}\beta^{AM}\beta^{M}} + {t_{RTX}\beta^{M}}} \right)}} \right\rbrack}}},{{{2 \times 10^{- 6}} < t_{1} < {{T_{1}\beta^{M}} + T_{{D\_ M} - {AM}}}};{t_{2} = {{t_{1} + {T_{1}\beta^{M}2 \times {\Delta\Phi}_{F_{1}/F_{2}}}} = {{\varphi_{BB\_ RECOV}^{M}\left( t_{2} \right)} - {\varphi_{BB\_ RECOV}^{M}\left( t_{1} \right)} - {\gamma^{M} \times \omega_{OSC} \times \left\lbrack \quad \right.K_{F_{2}}t_{2}} - {K_{F\; 1}t_{1}} - {\left( {K_{F_{2}} - K_{F_{1}}} \right) \times T_{1}\beta^{M}} - {\left( {K_{F_{2}} - K_{F_{1}}} \right) \times T_{LB\_ M}\beta^{M}} - {\left( {K_{F_{2}} - K_{F_{1}}} \right) \times \left( {{T_{LB\_ AM}\beta^{AM}\beta^{M}} + {t_{RTX}\beta^{M}}} \right)}}}}}} \right\rbrack,{{{2 \times 10^{- 6}} < t_{1} < {{T_{1}\beta^{M}} + T_{{D\_ M} - {AM}}}};{t_{2} = {t_{1} + {T_{1}\beta^{M}}}}}}{{{2 \times {\Delta\Phi}_{F_{1}/F_{2}}} = {{\varphi_{BB\_ RECOV}^{M}\left( t_{2} \right)} - {\varphi_{BB\_ RECOV}^{M}\left( t_{1} \right)} - {\gamma^{M} \times \omega_{OSC} \times \left\lbrack {{K_{F_{2}}t_{2}} - {K_{F\; 1}t_{1}} - {\left( {K_{F_{2}} - K_{F_{1}}} \right) \times \left( {{T_{1}\beta^{M}} - {T_{LB\_ M}\beta^{M}} - {T_{LB\_ AM}\beta^{AM}\beta^{M}} - {t_{RTX}\beta^{M}}} \right)}} \right\rbrack}}},{{{2 \times 10^{- 6}} < t_{1} < {{T_{1}\beta^{M}} + T_{{D\_ M} - {AM}}}};{t_{2} = {t_{1} + {T_{1}\beta^{M}}}};}}} & (6) \end{matrix}$ Or, assuming that β^(M)=β^(AM)=1: 2×ΔΦ_(F) ₁ _(/F) ₂ =φ_(BB) _(—) _(RECOV) ^(M)(t ₂)−φ_(BB) _(—) _(RECOV) ^(M)(t ₁)−γ^(M)×ω_(OSC) ×[K _(F) ₂ t ₂ −K _(F1) t ₁−(K _(F) ₂ −K _(F) ₁ )×(T ₁ −T _(D) _(—) _(M-AM))],  (6A)

From the equation (6) it can be concluded that at operating frequency(s) ranging signal(s) complex amplitude values can be found from processing the returned base-band ranging signal.

The initial phase value 2×φ_(F) ₁ ^(MULT) can be assumed to be equal zero because the subspace algorithms are not sensitive to a constant phase offset. If necessary, the 2×φ_(F) ₁ ^(MULT) value (phase initial value) can be found by determining the TOA (Time Of Arrival) using the narrow-bandwidth ranging signal method as described in a co-pending application Ser. No. 11/670,595, incorporated herein by reference in its entirety. This method estimates the ranging signal round trip delay, which is equal to 2×T_(FLT)β^(M) and the 2×φ_(F) ₁ ^(MULT) value can be found from the following equation: 2×φ_(F) ₁ ^(MULT)=2×β^(M)×γ^(M)×ω_(OSC)×(K _(SYN) _(—) _(TX) +K _(F) ₁ )×(T _(FLT)), Or: 2×φ_(F) ₁ ^(MULT)=2×ω_(OSC)×(K _(SYN) _(—) _(TX) +K _(F) ₁ )×(T _(FLT)),

In the preferred embodiment, the returned base-band ranging signal phase values φ_(BB) _(—) _(RECOV) ^(M)(t) are calculated by the multi-path processor' arctan block 250. To improve SNR, the multi-path mitigation processor phase compare block 255 calculates 2×ΔΦ_(F) ₁ _(F) _(i) =φ_(BB) _(—) _(RECOV) ^(M)(t_(m))−φ_(BB) _(—) _(RECOV) ^(M)(t_(n)) for many instances n (n=2, 3, 4 . . . ) using the equation (6A), and then average them out to improve SNR. Note that 2×10⁻⁶<t_(n)<T_(f)+T_(D) _(—) _(M-AM); t_(m)=t₁+T_(f).

From the equations 5 and 6 it becomes apparent that the recovered (i.e., received) base-band ranging signal has the same frequency as the original base-band signal that was sent by the master. Thus, there is no frequency translation despite the fact that the master (M) and the transponder (AM) system clocks can differ. Because the base-band signal consists of several frequency components, each component is consists of multiple periods of a sinusoid, it is also possible to estimate the phase and the amplitude of the received ranging signal by sampling the received base-band signal individual component frequency with the corresponding original (i.e., sent by the master) base-band signal individual frequency component and integrating the resulting signal over period T≦T_(f).

This operation generates complex amplitude values Â_(RT)(f_(n)) of received ranging signal in the I/Q format. Note that each base-band signal individual frequency component that was sent by the master has to be shifted in time by the T_(D) _(—) _(M-AM). The integration operation produces effect of averaging out the multiple instances of the amplitude and the phase (e.g., increasing the SNR). Note that the phase and the amplitude values can be translated from the I/Q format to the |Â(f_(n))| and ∠Â_(RT)f_(n)) format.

This method of sampling, integrating over period of T≦T_(f) and subsequent conversion from the I/Q format to the |Â(f_(n))| and ∠Â(f_(n)) format can be implemented in the phase compare block 255 in FIG. 3C. Thus, depending upon the block's 255 design and implementation, either the method of the preferred embodiment, based on the equation (5), or an alternative method, described in this section, can be used.

Although the ranging signal bandwidth is narrow, the frequency difference f_(n)−f₁ can be relatively large, for example, in an order of several megahertz. As a result, the receiver's bandwidth has to be kept wide enough to pass all of the f₁:f_(n) ranging signal frequencies components. This wide receiver bandwidth impacts the SNR. To reduce the receiver effective bandwidth and improve the SNR, the received ranging signal base-band frequency components can be filtered by the RF back-end processor in FPGA 150 by the digital narrow bandwidth filters tuned for each individual frequency component of the received base-band ranging signal. However, this large number of digital filters (the number of filters equals to the number of individual frequency components, n) puts additional burden on the FPGA resources, increasing its cost, size and power consumption.

In the preferred embodiment only two narrow bandwidth digital filters are used: one filter is always tuned for f₁ frequency component and the other filter can be tuned for all other frequencies components: f₂:f_(n). Multiple instances of ranging signal are sent by the master. Each instance consists of only two frequencies: f₁: f₂; f₁:f₃ . . . ; f₁:f_(i) . . . ; f₁:f_(n). Similar strategies are also possible.

Please note that it is also entirely possible to keep the base-band ranging signal components to only two (or even one) generating the rest of the frequency components by adjusting the frequency synthesizers, e.g. changing K_(SYN). It is desirable that LO signals for up-converters and down-converters mixers are generated using the Direct Digital Synthesis (DDS) technology. For high VHF band frequencies this can present an undesired burden on the transceiver/FPGA hardware. However, for lower frequencies this might be a useful approach. Analog frequency synthesizers can also be used, but may take additional time to settle after frequency is changed. Also, in case of analog synthesizers, two measurements at the same frequency would have to be made in order to cancel a phase offset that might develop after changing the analog synthesizer's frequency.

The actual T_(D) _(—) _(M-AM) that is used in the above equations is measured in both: the master (M) and the transponder (AM) systems clocks, e.g. T_(LB) _(—) _(AM) and t_(RTX) are counted in the transponder (AM) clocks and T_(LB) _(—) _(M) is counted in the master (M) clock. However, when 2×ΔΦ_(F) ₁ _(/F) ₂ is calculated both: T_(LB) _(—) _(AM) and t_(RTX) are measured (counted) in master (M) clock. This introduces an error: 2×ΔΦ_(ERROR)=Δ^(M)×ω_(OSC)×(K _(F) ₂ −K _(F) ₁ )×(T _(LB) _(—) _(AM)(β^(AM)β^(M)−β^(AM))+t _(RTX)(β^(M)−β^(AM)))  (7)

The phase estimation error (7) impacts the accuracy. Therefore, it is necessary to minimize this error. If β^(M)=β^(AM), in other words, all master(s) and transponders (tags) system clocks are synchronized, then the contribution from the t_(RTX) time is eliminated.

In the preferred embodiment, the master and the transponder units (devices) are capable of synchronizing clocks with any of the devices. For example, a master device can serve as a reference. Clock synchronization is accomplished by using the remote control communication channel, whereby under FPGA 150 control, the frequency of temperature compensated crystal oscillator TCXO 20 is adjusted. The frequency difference is measured at the output of the summer 270 of the master device while the selected transponder device is transmitting a carrier signal.

Thereafter, the master sends a command to the transponder to increase/decrease TCXO frequency. This procedure may be repeated several times to achieve greater accuracy by minimizing frequency at the summer 270 output. Please note that in an ideal case the frequency at the summer 270 output should become equal to zero. An alternative method is to measure the frequency difference and make a correction of the estimated phase without adjusting the transponder' TCXO frequency.

While β^(M)−β^(AM) can be considerably reduced there is a phase estimation error when β^(M)≠1. In this case the margin of error depends upon a long term stability of the reference device (usually master (M)) clock generator. In addition, the process of clock synchronization may take considerable amount of time, especially with large number of units in the field. During the synchronization process the track-locate system becomes partially or fully inoperable, which negatively impacts the system readiness and performance. In this case the abovementioned method that does not require the transponder' TCXO frequency adjustment is preferred.

Commercially available (off the shelf) TCXO components have high degree of accuracy and stability. Specifically, TCXO components for the GPS commercial applications are very accurate. With these devices, the phase error impact on locating accuracy can be less than one meter without the need for frequent clock synchronization.

After narrow bandwidth ranging signal multi-path mitigation processor obtains the returned narrow bandwidth ranging signal complex amplitude Â_(RT)(f_(n)), the further processing (i.e., execution of super-resolution algorithms), is implemented in the software-based component, which is a part of the multi-path mitigation processor. This software component can be implemented in the master (reader) host computer CPU and/or the microprocessor that is embedded in the FPGA 150 (not shown). In the preferred embodiment the multi-path mitigation algorithm(s) software component is executed by the master host computer CPU.

The super-resolution algorithm(s) produce estimation of (2π×τ_(K)) “frequencies”, e.g. τ_(K) values. At the final step the multi-path mitigation processor selects τ with the smallest value (i.e., the DLOS delay time).

In certain cases where the ranging signal narrow bandwidth requirements are somewhat relaxed, the DLOS path can be separated from MP paths by employing a continuous (in time) chirp. In the preferred embodiment this continuous chirp is Linear Frequency Modulation (LFM). However, other chirp waveforms can be also used.

Let's assume that under multi-path mitigation processor control a chirp with bandwidth of B and duration of T is transmitted. That gives a chirp rate of

$\beta = {2\pi\frac{B}{T}}$ radians per second. Multiple chirps are transmitted and received back. Note that chirps signals are generated digitally with each chirp started at the same phase.

In the multi-path processor each received single chirp is aligned so that the returned chirp is from the middle of the area of interest.

The chirp waveform equation is: s(t)=exp(i(ω₀ t+βt ²)), where ω₀ is the initial frequency for 0<t<T. For a single delay round-trip τ, e.g. no multi-path, the returned signal (cirp) is s(t−τ).

The multi-path mitigation processor then “deramps” the s(t−τ) by performing complex conjugate mix with the originally transmitted chirp. The resulting signal is a complex sinusoid: f _(τ)(t)=exp(−ω₀τ)exp(−2iβτt)exp(iβτ ²),  (8)

-   -   where exp(−iw₀τ_(k)) is the amplitude and 2βτ is the frequency         and 0≦t≦T. Note that the last term is a phase and it is         negligible.

In case of multi-path, the composite deramped signal consists of multiple complex sinusoids:

$\begin{matrix} {{{f_{MP}(t)} = {\sum\limits_{k = 0}^{k = L}\;{{\exp\left( {{- {\mathbb{i}}}\; w_{0}\tau_{k}} \right)}{\exp\left( {- {\mathbb{i}2\beta\tau}_{k}} \right)}(t)}}},} & (9) \end{matrix}$

where L is the number of ranging signal paths, including the DLOS path and 0≦t≦T.

Multiple chirps are transmitted and processed. Each chirp is individually treated/processed as described above. Thereafter, the multi-path mitigation processor assembles results of individual chirps processing:

$\begin{matrix} {{f_{MP}^{N}(t)} = {\left\lbrack {\sum\limits_{n = 0}^{n = {N - 1}}\;{P\left( {t - {n\;\rho}} \right)}} \right\rbrack \times \left\lbrack {\sum\limits_{k = 0}^{k = L}\;{{\exp\left( {{- {\mathbb{i}}}\; w_{0}\tau_{k}} \right)}{\exp\left( {- {\mathbb{i}2\beta\tau}_{k}} \right)}t}} \right\rbrack}} & (10) \end{matrix}$

where N is the number of chirps,

${{P(t)} = \begin{Bmatrix} {1;{0 \leq t \leq T}} \\ {0;{t > T}} \end{Bmatrix}},$ ρ=T+t_(dead); t_(dead) is the dead time zone between two consecutive chirps; 2βτ_(k) are artificial delay “frequencies”. Again, the most interesting is the lowest “frequency”, which corresponds to the DLOS path delay.

In the equation (10) f_(MP) ^(N)(t) can be thought of as N samples of a sum of complex sinusoids at times: 0≦t _(α) ≦T;t ₁ =t _(α) +ρ;t ₂ =t _(α)+2ρ . . . ;t _(m-1) =t _(α)+(N−1)ρ;mε0;m−1 Thus, the number of samples can be a multiple of N, e.g. αN; α=1, 2, . . . .

From the equation (10) the multi-path mitigation processor produces αN complex amplitude samples in time domain that are used in further processing (i.e., execution of super-resolution algorithms). This further processing is implemented in the software component, which is a part of the multi-path mitigation processor. This software component can be executed by the master (reader) host computer CPU and/or by the microprocessor that is embedded in the FPGA 150 (not shown), or both. In the preferred embodiment the multi-path mitigation algorithm(s) software is executed by the master host computer CPU.

The super-resolution algorithm(s) produce estimation of 2βτ_(k) “frequencies”, e.g. τ_(K) values. At the final step the multi-path mitigation processor selects τ with the smallest value, i.e. the DLOS delay time.

An explanation will be given of a special processing method, called the “threshold technique,” which can serve as an alternative to the super-resolution algorithms. In other words, it is used to enhance reliability and accuracy in distinguishing DLOS path from other MP paths using the artificially generated synthetic wider bandwidth ranging signal.

The frequency domain base-band ranging signal shown in FIG. 1 and FIG. 1A can be converted into time domain base-band signal s(t):

$\begin{matrix} {{s(t)} = \frac{\sin\mspace{14mu}{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}}{\sin\mspace{14mu}{{\pi\Delta}{ft}}}} & (11) \end{matrix}$ It is readily verified that s(t) is periodic with period 1/Δt, and for any integer k, that s(k/Δt)=2N+1, which is the peak value of the signal. Where n=N in FIG. 1 and FIG. 1A.

FIG. 4 shows two periods of s(t) for the case where N=11 and Δf=250 kHz. The signal appears as a sequence of pulses of height 2N+1=23 separated by 1/Δf=4 microseconds. Between the pulses is a sinusoidal waveform with varying amplitude and 2N zeros. The wide bandwidth of the signal can be attributed to the narrowness of the tall pulses. It can be also seen that the bandwidth extends from zero frequency to NΔf=2.75 MHz.

The basic idea of the thresholded method that is used in the preferred embodiment is to enhance the artificially generated synthetic wider bandwidth ranging reliability and accuracy in distinguishing DLOS path from other MP paths. The threshold method detects when the start of the leading edge of a wideband pulse arrives at a receiver. Because of filtering in the transmitter and receiver, the leading edge does not rise instantaneously, but rises out of the noise with smoothly increasing slope. The TOA of the leading edge is measured by detecting when the leading edge crosses a predetermined threshold T.

A small threshold is desirable because it gets crossed sooner and the error delay τ between the true start of the pulse and the threshold crossing is small. Thus, any pulse replica arriving due to multi-path has no effect if the start of the replica having a delay greater than τ. However, the presence of noise places a limit on how small the threshold T can be. One way to decrease the delay τ is to use the derivative of the received pulse instead of the pulse itself, because the derivative rises faster. The second derivative has an even faster rise. Higher order derivatives might be used, but in practice they can raise the noise level to an unacceptable value, so the thresholded second derivative is used.

Although the 2.75 MHz wide signal depicted in FIG. 4 has a fairly wide bandwidth, it is not suitable for measuring range by the abovementioned method. That method requires transmitted pulses each having a zero-signal precursor. However, it is possible to achieve that goal by modifying the signal so that the sinusoidal waveform between the pulses is essentially cancelled out. In the preferred embodiment it is done by constructing a waveform which closely approximates the signal on a chosen interval between the tall pulses, and then subtracting it from the original signal.

The technique can be illustrated by applying it to the signal in FIG. 1. The two black dots shown on the waveform are the endpoints of an interval I centered between the first two pulses. The left and right endpoints of the interval I, which have been experimentally determined to provide the best results, are respectively at:

$\begin{matrix} \begin{matrix} {t_{1} = {\frac{1.1}{\left( {{2N} + 1} \right)\Delta\; f} = {\frac{1.1}{23 \times 250\text{,}000} \cong {191.3\mspace{14mu}{nsec}}}}} \\ {t_{2} = {{\frac{1}{\Delta\; f} - t_{1}} = {{\frac{1}{250\text{,}000} - \frac{1.1}{23 \times 250\text{,}000}} \cong {3\text{,}808.7\mspace{14mu}{nsec}}}}} \end{matrix} & (12) \end{matrix}$

An attempt to generate a function g(t) which essentially cancels out the signal s(t) on this interval, but does not cause much harm outside the interval, is performed. Since the expression (11) indicates that s(t) is the sinusoid sin π(2N+1)Δft modulated by 1/sin πΔft, first a function h(t) which closely approximates 1/sin πΔft on the interval I is found, and then form g (t) as the product: g(t)=h(t)sin π(2N+1)Δft  (13) h(t) is generated by the following sum:

$\begin{matrix} {{{h(t)} = {\sum\limits_{k = 0}^{M}\;{a_{k}{\phi_{k}(t)}{dt}}}},\mspace{14mu}{t \in I}} & (14) \end{matrix}$ where φ₀(t)≡1, φ_(k)(t)=sin kπΔft for k=1,2, . . . ,M  (15) and the coefficients α_(k) are chosen to minimize the least-square error

$\begin{matrix} {J = {\int_{t_{1}}^{t_{2}}{\left( {{{1/\sin}\mspace{14mu}{{\pi\Delta}{ft}}} - {\sum\limits_{k = 0}^{M}\;{a_{k}{\phi_{k}(t)}}}} \right)^{2}\ {\mathbb{d}t}}}} & (16) \end{matrix}$ over the interval I.

The solution is readily obtained by taking partial derivatives of J with respect to the α_(k) and setting them equal to zero. The result is the linear system of M+1 equations

$\begin{matrix} {{{\sum\limits_{k = 0}^{M}\;{a_{k}R_{jk}}} = R_{j}},\mspace{14mu}{j = 0},1,2,\ldots\mspace{14mu},M} & (17) \end{matrix}$

that can be solved for the α_(k), where

$\begin{matrix} {{R_{j} = {\int_{t_{1}}^{t_{2}}{{\phi_{j} \cdot {1/\sin}}\mspace{14mu}{{\pi\Delta}{ft}}\ {\mathbb{d}t}}}},\mspace{14mu}{R_{jk} = {\int_{t_{1}}^{t_{2}}{{\phi_{j}(t)}{\phi_{k}(t)}\ {\mathbb{d}t}}}}} & (18) \end{matrix}$

Then,

$\quad\begin{matrix} \begin{matrix} {{g(t)} = {{h(t)}\sin\mspace{14mu}{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}}} \\ {= {\left( {\sum\limits_{k = 0}^{M}\;{a_{k}{\phi_{k}(t)}}} \right)\sin\mspace{14mu}{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}}} \end{matrix} & (19) \end{matrix}$

Using the definition of the functions φ_(k)(t) given by (12)

$\begin{matrix} {{g(t)} = {\left( {a_{0} + {\sum\limits_{k = 1}^{M}\;{a_{k}\sin\mspace{11mu} k\;{{\pi\Delta}{ft}}}}} \right)\sin\mspace{11mu}{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}}} & (20) \end{matrix}$

The g(t) is subtracted from s(t) to get a function r(t), which should essentially cancel s(t) on the interval I. As indicated in the Appendix, an appropriate choice for the upper limit M for the summation in the equation (20) is M=2N+1. Using this value and the results from the Appendix,

$\begin{matrix} \begin{matrix} {{r(t)} = {{s(t)} - {g(t)}}} \\ {= {b_{0} + {\sum\limits_{k = 1}^{{2N} + 1}\;{b_{k}\cos\; 2\pi\; k\;{\Delta{ft}}}} + {c\mspace{11mu}\sin\mspace{11mu} 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}}}} \end{matrix} & (21) \end{matrix}$ where

$\begin{matrix} \begin{matrix} {b_{0} = {1 - {\frac{1}{2}a_{{2N} + 1}}}} \\ {{b_{k} = {{2 - {\frac{1}{2}a_{{2{({N - k})}} + 1}\mspace{14mu}{for}\mspace{14mu} k}} = 1}},2,\ldots\mspace{14mu},N} \\ {{b_{k} = {{{- \frac{1}{2}}a_{{2{({k - N})}} - 1}\mspace{14mu}{for}\mspace{14mu} k} = {N + 1}}},{N + 2},\ldots\mspace{14mu},{{2N} + 1}} \\ {c = {- a_{0}}} \end{matrix} & (22) \end{matrix}$

From the equation (17) it is seen that a total of 2N+3 frequencies (including the zero-frequency DC term) are required to obtain the desired signal r(t). FIG. 5 shows the resulting signal r(t) for the original signal s(t) shown in FIG. 1, where N=11. In this case the construction of r(t) requires 25 carriers (including the DC term b₀).

The important characteristics of r(t) as constructed above are as follows:

1. The lowest frequency is zero Hz and the highest frequency is (2N+1)Δf Hz, as seen from (14). Thus, the total bandwidth is (2N+1)Δf Hz.

2. All carriers are cosine functions (including DC) spaced Δf apart, except for one carrier, which is a sine function located at frequency

$\left( {N + \frac{1}{2}} \right)\Delta\;{f.}$

3. Although the original signal s(t) has period 1/Δf, r(t) has period 2/Δf. The first half of each period of r(t), which is a full period of s(t), contains a cancelled portion of the signal, and the second half-period of r(t) is a large oscillatory segment. Thus, cancellation of the precursor occurs only in every other period of s(t).

This occurs because the canceling function g(t) actually strengthens s(t) in every other period of s(t). The reason is that g (t) reverses its polarity at every peak of s(t), whereas s(t) does not. A method of making every period of s(t) contain a cancelled portion to increase processing gain by 3 dB is described below.

4. The length of the cancelled portion of s(t) is about 80-90% of 1/Δf. Therefore, Δf needs to be small enough to make this length long enough to eliminate any residual signal from previous non-zero portions of r(t) due to multi-path.

5. Immediately following each zero portion of r(t) is the first cycle of an oscillatory portion. In the preferred embodiment, in the TOA measurement method as described above, the first half of this cycle is used for measuring TOA, specifically the beginning of its rise. It is interesting to note that the peak value of this first half-cycle (which will be called the main peak) is somewhat larger than the corresponding peak of s(t) located at approximately the same point in time. The width of the first half-cycle is roughly inversely proportional to NΔf.

6. A large amount of processing gain can be achieved by:

(a) Using the repetitions of the signal r(t), because r(t) is periodic with period 2/Δf. Also, an additional 3 dB of processing gain is possible by a method to be described later.

(b) Narrowband filtering. Because each of the 2N+3 carriers is a narrowband signal, the occupied bandwidth of the signal is much smaller than that of a wideband signal spread out across the entire allocated band of frequencies.

For the signal r(t) shown in FIG. 5, where N=11 and Δf=250 kHz, the length of the cancelled portion of s(t) is about 3.7 microseconds or 1,110 meters. This is more than enough to eliminate any residual signal from previous non-zero portions of r(t) due to the multi-path. The main peak has value of approximately 35, and the largest magnitude in the precursor (i.e., cancellation) region is about 0.02, which is 65 dB below the main peak. This is desirable for getting good performance using the TOA measurement thresholded technique as described above.

Use of fewer carriers is depicted in FIG. 6, which illustrates a signal that is generated using Δf=850 kHz, N=3, and M=2N+1=7, for a total of only 2N+3=9 carriers. In this case, the period of the signal is only 2/Δf≅2.35 microseconds as compared to the signal in FIG. 5, where the period is 8 microseconds. Since this example has more periods per unit time, one might expect that more processing gain could be achieved.

However, since fewer carriers are used, the amplitude of the main peak is about ⅓ as large as before, which tends to cancel the expected extra processing gain. Also, the length of the zero-signal precursor segments is shorter, about 0.8 microseconds or 240 meters. This should still be enough to eliminate any residual signal from previous non-zero portions of r(t) due to the multi-path. Note that the total bandwidth of (2N+1)Δf=5.95 MHz is about the same as before, and that the width of the half-cycle of the main peak is also roughly the same. Since fewer carriers are used, there should be some extra processing gain when each carrier is narrowband filtered at the receiver. Moreover, the largest magnitude in the precursor (i.e., cancellation) region is now about 75 dB below the main peak, a 10 dB improvement from the previous example.

Transmission at RF Frequencies: up to this point r(t) has been described as a base-band signal for purposes of simplicity. However, it can be translated up to RF, transmitted, received, and then reconstituted as a base-band signal at the receiver. To illustrate, consider what happens to one of the frequency components ω_(k) in the base-band signal r(t) traveling via one of the multi-path propagation paths having index j (radian/sec frequencies are used for notational simplicity): b_(k) cos ω_(k)t (at baseband in transmitter) b_(k) cos(ω+ω_(k))t (translated by frequency ωup to RF) a_(j)b_(k) cos [(ω+ω_(k))(t−τ_(j))+φ_(j)] (at receiver antenna) a_(j)b_(k) cos [ω_(k)(t−τ_(j))+φ_(j)+θ] (translated by frequency −ω to baseband)  (23)

It is assumed here that the transmitter and receiver are frequency synchronized. The parameter b_(k) is the k^(th) coefficient in expression (21) for r(t). The parameters τ_(j) and φ_(j) are respectively the path delay and phase shift (due to dielectric properties of a reflector) of the j^(th) propagation path. The parameter θ is the phase shift occurring in the down-conversion to base-band in the receiver. A similar sequence of functions can be presented for the sine component of the equation (21).

It is important to note that as long as the zero-signal precursors in r(t) have length sufficiently larger than the largest significant propagation delay, the final base-band signal in the equation (20) will still have zero-signal precursors. Of course, when all frequency components (index k) over all paths (index j) are combined, the base-band signal at the receiver will be a distorted version of r (t), including all phase shifts.

Sequential Carrier Transmissions and Signal Reconstruction are illustrated in FIG. 1 and FIG. 1A. It is assumed that the transmitter and the receiver are time and frequency synchronized, the 2N+3 transmitted carriers need not be transmitted simultaneously. As an example, consider the transmission of the signal whose base-band representation is that of FIG. 1A and FIG. 6.

In FIG. 6 N=3, and suppose each of the 9 frequency components for 1 millisecond are sequentially transmitted. The start and the end times for each frequency transmission are known at the receiver, so it can sequentially start and end its reception of each frequency component at those respective times. Since the signal propagation time is very short compared to 1 millisecond (it will normally be less than several microseconds in the intended application), a small portion of each received frequency component should be ignored, and the receiver can easily blank it out.

The entire process of receiving 9 frequency components can be repeated in 9-millisecond blocks of additional reception to increase the processing gain. In one second of total reception time there would be about 111 such 9-millisecond blocks available for processing gain. Additionally, within each block there would be additional processing gain available from 0.009/(2/Δf)≅383 main peaks.

It is worth noting that in general the signal reconstruction can be made very economical, and will inherently permit all possible processing gain. For each of the 2N+3 received frequencies:

1. Measure the phase and amplitude of each 1-millisecond reception of that frequency to form a sequence of stored vectors (phasors) corresponding to that frequency.

2. Average the stored vectors for that frequency.

3. Finally, use the 2N+3 vector averages for the 2N+3 frequencies to reconstruct 1 period of base-band signal having duration 2/Δf, and use the reconstruction to estimate signal TOA.

This method is not restricted to 1-millisecond transmissions, and the length of the transmissions may be increased or decreased. However, the total time for all transmissions should be short enough to freeze any motion of the receiver or transmitter.

Obtaining Cancellation on Alternate Half-Cycles of r(t): by simply reversing the polarity of the canceling function g(t), cancellation between the peaks of s(t) is possible where r(t) was formerly oscillatory. However, to obtain cancellation between all peaks of s(t), the function g(t) and its polarity reversed version must be applied at the receiver, and this involves coefficient weighting at the receiver.

Coefficient Weighting at the Receiver: if desired, the coefficients b_(k) in the equation (21) are used for construction of r(t) at the transmitter and may be introduced at the receiver instead. This is easily seen by considering the sequence of signals in the equation (20) in which the final signal is the same if b_(k) is introduced at the last step instead of at the beginning. Ignoring noise, the values are as follows: cos ω_(k)t (at baseband in transmitter) cos(ω+ω_(k))t (translated by frequency ω up to RF) a_(j) cos [(ω+ω_(k))(t−τ_(j))+φ_(j)] (at receiver antenna) a _(j) cos [ω_(k) (t−τ _(j))+φ_(j)+θ] (translated by frequency −ω to baseband) a_(j)b_(k) cos [ω_(k)(t−τ_(j))+φ_(j)+θ] (weighted by coefficient b_(k) at baseband)  (24)

The transmitter can then transmit all frequencies with the same amplitude, which simplifies its design. It should be noted, that this method also weights the noise at each frequency, the effect of which should be considered. It should also be noted that coefficient weighting should be done at the receiver in order to effect the polarity reversal of g(t) to get twice as many useable main peaks.

Scaling of Δf to Center Frequencies in Channels: to meet the FCC requirements at the VHF or lower frequencies a channelized transmission with constant channel spacing will be required. In a channelized transmission band with constant channel spacing that is small compared to the total allocated band, which is the case for the VHF and lower frequencies band(s), small adjustments to Δf, if necessary, permit all transmitted frequencies to be at channel centers without materially changing performance from original design values. In the two examples of base-band signals previously presented, all frequency components are multiples of Δf/2, so if the channel spacing divides Δf/2, the lowest RF transmitted frequency can be centered in one channel and all other frequencies fall at the center of channels.

In some Radio Frequency (RF)-based identification, tracking and locating systems in addition to performing the distance measurement function, both: the Master Unit and the Tag Unit also perform voice, data and control communication functions. Similarly, in the preferred embodiment both the Master Unit and the Tag perform voice, data and control communication functions in addition to the distance measurement function.

According to the preferred embodiment, the ranging signal(s) are subject to the extensive sophisticated signal processing techniques, including the multi-path mitigation. However, these techniques may not lend themselves to the voice, data and control signals. As a result, the operating range of the proposed system (as well as other existing systems) may be limited not by its ability to measure distance reliably and accurately, but by being out of range during voice and/or data and/or control communications.

In other Radio Frequency (RF)-based identification, tracking and locating systems the distance measurement functionality is separated from the voice, data and control communication functionality. In these systems separate RF Transceivers are used to perform voice, data and control communication functions. The drawback of this approach is system increased cost, complexity, size, etc.

To avoid abovementioned drawbacks, in the preferred embodiment, a narrow bandwidth ranging signal or base-band narrow bandwidth ranging signal several individual frequency components are modulated with the identical data/control signals and in case of voice with digitized voice packets data. At the receiver the individual frequency components that have the highest signal strength are demodulated and the obtained information reliability may be further enhanced by performing “voting” or other signal processing techniques that utilize the information redundancy.

This method allows to avoid the “null” phenomena, wherein the incoming RF signals from multiple paths are destructively combining with the DLOS path and each other, thus significantly reducing the received signal strength and associated with it SNR. Moreover, such method allows to find a set of frequencies at which the incoming signals from multiple paths are constructively combining with DLOS path and each other, thus increasing the received signal strength and associated with it SNR.

As mentioned earlier, spectrum estimation-based super-resolution algorithms generally use the same model: a linear combination of complex exponentials and their complex amplitudes of frequencies. This complex amplitude is given by equation 3 above.

All spectrum estimation-based super-resolution algorithms require a priori knowledge of number of complex exponentials, i.e., the number of multipath paths. This number of complex exponentials is called the model size and is determined by the number of multi-path components L as shown in equations 1-3. However, when estimating path delay, which is the case for RF track-locate applications, this information is not available. This adds another dimension, i.e., the model size estimation, to the spectrum estimation process via super-resolution algorithms.

It has been shown (Kei Sakaguchi et al., Influence of the Model Order Estimation Error in the ESPRIT Based High Resolution Techniques) that in case of model size underestimation the accuracy of frequency estimation is impacted and when the model size is overestimated the algorithm generates spurious, e.g., non-existent, frequencies. Existing methods of model size estimation such as AIC (Akaikes Information Criterion), MDL (Minimum Description Length), etc. have a high sensitivity to correlation between signals (complex exponentials). But in the case of RF multipath, this is always the case. Even, for example, after Forward-Backward smoothing algorithms are applied, there will always be a residual amount of correlation.

In the Sakaguchi paper, it is suggested to use an overestimated model and differentiating actual frequencies (signals) from spurious frequencies (signals) by estimating these signals power (amplitude) and then rejecting the signals with very low power. Although this method is an improvement over existing methods, it is not guaranteed. The inventors implemented the Kei Sakaguchi et al. method and ran simulations for more complex cases with a larger model size. It was observed that, in some cases, a spurious signal may have amplitude that is very close to actual signals amplitude.

All spectrum estimation-based super-resolution algorithms work by splitting the incoming signal complex amplitude data into two sub-spaces: the noise sub-space and signals sub-space. If these sub-spaces are properly defined (separated), then the model size is equal to the signal sub-space size (dimension).

In one embodiment of the present invention, the model size estimation is accomplished using an “F” statistic. For example, for ESPRIT algorithm, the singular value decomposition of the estimate of the covariance matrix (with forward/backward correlation smoothing) is ordered in ascending order. Thereafter, a division is made whereby the (n+1) eigenvalue is divided by the n-th eigenvalue. This ratio is an “F” random variable. The worst case is an “F” random variable of (1,1) degree of freedom. The 95% confidence interval for a “F” random variable with (1,1) degrees of freedom is 161. Setting that value as a threshold determines the model size. Note also that for the noise subspace, the eigenvalues represent an estimate of the noise power.

This method of applying “F” statistics to the ratio of the eigenvalues is a more accurate method of estimating the model size. It should be noted that other degrees of freedom in “F” statistics can be also used for threshold calculation and consequently model size estimation.

Nevertheless, in some cases, two or more very closely spaced (in time) signals can degenerate into one signal because of real-world measurement imperfections. As a result, the above mentioned method will underestimate the number of signals, i.e., the model size. Since model size underestimation reduces the frequency estimation accuracy, it is prudent to increase the model size by adding a certain number. This number can be determined experimentally and/or from simulations. However, when signals are not closely spaced, the model size will be overestimated.

In such cases spurious, i.e., non-existent, frequencies may appear. As noted earlier, using signal amplitude for spurious signals detection does not always work because in some cases a spurious signal(s) was observed to have amplitude that is very close to actual signal(s) amplitude. Therefore, in addition to the amplitude discrimination, filters can be implemented to improve spurious frequencies elimination probability.

The frequencies that are estimated by super-resolution algorithms are artificial frequencies (equation 2). In fact, these frequencies are individual paths delays of the multipath environment. As a result, there should be no negative frequencies and all negative frequencies that are produced by a super-resolution algorithm are spurious frequencies to be rejected.

Furthermore, a DLOS distance range can be estimated from the complex amplitude Â(f_(n)) values obtained during measurements using methods that are different from super-resolution methods. While these methods have lower accuracy, this approach establishes range that is used to discriminate delays, i.e., frequencies. For example, the ratio of

$\frac{\Delta\left\lbrack {\angle{\hat{A}\left( {2{\pi\Delta}\; f} \right)}} \right\rbrack}{2{\pi\Delta}\; f}$

in Δf intervals where the signal amplitude |Â(f_(n))| is close to maximum, i.e., avoiding nulls, provides a DLOS delay range. Although actual DLOS delay can be up to two times larger or smaller, this defines a range that helps to reject spurious results.

In the exemplary embodiment, the ranging signal makes a round-trip. In other words, it travels both ways: from a master/reader to a target/slave and from the target/slave back to the master/reader:

Master transmits a tone: α×e^(−jωt), where ω is an operating frequency in the operating band and α is the tone signal amplitude.

At the target's receiver, the received signal (one-way) is as follows:

$\begin{matrix} {{S_{{one} - {way}}(t)} = {\alpha \times {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{{- {j\omega}}\; t} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}}}} & (25) \end{matrix}$

Where: N is number of signal paths in the multipath environment; K0 and τ₀ are amplitude and time-of-flight of the DLOS signal; |K₀|=1, K₀>0, |K_(m≠0)|≦1 and K_(m≠0) can be positive or negative. S _(one-way)(t)=α×e ^(−jωt) ×A(ω)×e ^(−jθ(ω))  (26)

Where:

${{A(\omega)} \times {\mathbb{e}}^{- {{j\theta}{(\omega)}}}} = {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}}$ is one way multipath RF channel transfer function in the frequency domain; and A(ω)≧0.

Target retransmits the received signal: S _(retransmit)(t)=α×e ^(−jωt) ×A(ω)×e ^(−jθ(ω))  (27)

At the master receiver, the round-trip signal is:

${S_{round\_ trip}(t)} = {\alpha \times {\mathbb{e}}^{{- {j\omega}}\; t} \times {A(\omega)} \times {\mathbb{e}}^{- {{j\theta}{(\omega)}}} \times {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}}}$

Or: S _(round) _(—) _(trip)(t)=α×e ^(−jωt) ×A ²(ω)×e ^(−j2θ(ω))  (28)

On the other hand from equations (26) and (28):

$\begin{matrix} {{S_{round\_ trip}(t)} = {\alpha \times {\mathbb{e}}^{{- {j\omega}}\; t} \times {A^{2}(\omega)} \times \left( {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}} \right)^{2}}} & (29) \end{matrix}$

Where:

${{A^{2}(\omega)} \times \left( {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}} \right)^{2}} = {{A^{2}(\omega)} \times {\mathbb{e}}^{- {{j2\theta}{(\omega)}}}}$ is roundtrip multipath RF channel transfer function in the frequency domain.

From equation 29, the roundtrip multipath channel has a larger number of paths than

one-way channel multipath because the

$\left( {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}} \right)^{2}$ expression in addition to the τ₀÷τ_(N) paths delays, includes combinations of these paths delays, for example: τ₀+τ₁, τ₀+τ₂ . . . , τ₁+τ₂, τ₁+τ₃, . . . , etc.

These combinations dramatically increase the number of signals (complex exponentials). Hence the probability of very closely spaced (in time) signals will also increase and may lead to significant model size underestimation. Thus, it is desirable to obtain one-way multipath RF channel transfer function.

In preferred embodiment, the one-way amplitude values |Â(f_(n))| are directly obtainable from target/slave device. However, the one-way phase values ∠Â(f_(n)) cannot be measured directly. It is possible to determine the phase of the one-way from the roundtrip phase measurements observation:

$\left( {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}} \right)^{2} = {{{\mathbb{e}}^{- {{j2\theta}{(\omega)}}}\mspace{14mu}{and}\mspace{14mu}\left( {\sum\limits_{m = 0}^{m = N}\;{K_{m} \times {\mathbb{e}}^{- {j\omega\tau}_{m}}}} \right)} = {\mathbb{e}}^{- {{j\theta}{(\omega)}}}}$

However, for each value of ω, there are two values of phase α(ω) such that e^(j2α(ω))=e^(jβ(ω))

A detailed description of resolving this ambiguity is shown below. If the ranging signal different frequency components are close to each other, then for most part the one-way phase can be found by dividing the roundtrip phase by two. Exceptions will include the areas that are close to the “null”, where the phase can undergo a significant change even with small frequency step. Note: the “null” phenomena is where the incoming RF signals from multiple paths are destructively combining with the DLOS path and each other, thus significantly reducing the received signal strength and associated with it SNR.

Let h(t) be the one-way impulse response of a communications channel. The corresponding transfer function in the frequency domain is

$\begin{matrix} {{H(\omega)} = {{\int_{- \infty}^{\infty}{{h(t)}{\mathbb{e}}^{{- {j\omega}}\; t}\ {\mathbb{d}t}}} = {{A(\omega)}{\mathbb{e}}^{{j\alpha}{(\omega)}}}}} & (30) \end{matrix}$ where A(ω)≧0 is the magnitude and α(ω) is the phase of the transfer function. If the one-way impulse response is retransmitted back through the same channel as it is being received, the resulting two-way transfer function is G(ω)=B(ω)e ^(jβ(ω)) =H ²(ω)=A ²(ω)e ^(j2α(ω))  (31)

where B(ω)≧0. Suppose the two-way transfer function G(ω) is known for all ω in some open frequency interval (ω₁, ω₂). Is it possible to determine the one-way transfer function H(ω) defined on (ω₁, ω₂) that produced G(ω)?

Since the magnitude of the two-way transfer function is the square of the one-way magnitude, it is clear that A(ω)=√{square root over (B(ω))}  (32)

However, in trying to recover the phase of the one-way transfer function from observation of G(ω), the situation is more subtle. For each value of ω, there are two values of phase α(ω) such that e^(j2α(ω))=e^(jβ(ω))  (33)

A large number of different solutions might be generated by independently choosing one of two possible phase values for each different frequency ω.

The following theorems, which assume that any one-way transfer function is continuous at all frequencies, help resolve this situation.

Theorem 1: Let I be an open interval of frequencies w containing no zeros of the two-way transfer function G(ω)=B (ω)e^(jβ(ω)). Let J(ω)=√{square root over (B(ω)e^(jγ(ω)))}{square root over (B(ω)e^(jγ(ω)))} be a continuous function on I where β(ω)=2γ(ω). Then J(ω) and −J(ω) are the one-way transfer functions which produce G(ω) on I, and there are no others.

Proof: One of the solutions for the one-way transfer function is the function H(ω)=√{square root over (B(ω))}e^(jα(ω)), continuous on I since it is differentiable on I, and where β(ω)=2α(ω). Since G(ω)≠0 on I, H(ω) and J(ω) are nonzero on I. Then,

$\begin{matrix} {\frac{H(\omega)}{J(\omega)} = {\frac{\sqrt{B(\omega)}{\mathbb{e}}^{{j\alpha}{(\omega)}}}{\sqrt{B(\omega)}{\mathbb{e}}^{{j\gamma}{(\omega)}}} = {\mathbb{e}}^{j{\lbrack{{\alpha{(\omega)}} - {\gamma{(\omega)}}}\rbrack}}}} & (34) \end{matrix}$

Since H(ω) and J(ω) are continuous and nonzero on I, their ratio is continuous on I, hence the right side of (34) is continuous on I. The conditions β(ω)=2α(ω)=2γ(ω) imply that for each ωεI, α(ω)−γ(ω) is either 0 or π. However, α(ω)−γ(ω) cannot switch between these two values without causing a discontinuity on the right side of (34). Thus, either α(ω)−γ(ω)=0 for all ωεI, or α(ω)−γ(ω)=π for all ωεI. In the first case, we get J(ω)=H(ω), and in the second we get J(ω)=−H(ω).

This theorem proves that to get a one-way solution on any open interval I containing no zeros of the transfer function G(ω)=B(ω)e^(jβ(ω)), we form the function J(ω)=√{square root over (B(ω))}e^(jγ(ω)), choosing the values of γ(ω) satisfying β(ω)=2γ(ω) in such a way as to make J(ω) continuous. Since it is known that there is a solution having this property, namely H(ω), it is always possible to do this.

An alternate procedure for finding a one-way solution is based on the following theorem:

Theorem 2: Let H(ω)=A(ω)e^(jα(ω)) be a one-way transfer function and let I be an open interval of frequencies w containing no zeros of H(ω). Then the phase function α(ω) of H(ω) must be continuous on I.

Proof: Let ω₀ be a frequency in the interval I. In FIG. 7, the complex value H(ω₀) has been plotted as a point in the complex plane, and by hypothesis, H(ω₀)≠0. Let ε>0 be an arbitrarily small real number, and consider the two angles of measure ε shown in the FIG. 7, as well as the circle centered at H(ω₀) and tangent to the two rays OA and OB. By assumption, H(ω) is continuous for all ω. Thus, if ω is sufficiently close to ω₀, the complex value H(ω) will lie in the circle, and it is seen that |α(ω)−α(ω₀|<ε. Since ε>0 was chosen arbitrarily, we conclude that α(ω)→α(ω₀) as ω→ω₀, so that the phase function α(ω) is continuous at ω₀.

Theorem 3: Let I be an open interval of frequencies ω containing no zeros of the two-way transfer function G(ω)=B (ω)e^(jβ(ω)). Let J(ω)=√{square root over (B(ω))}e^(jγ(ω)) be a function on I where β(ω)=2γ(ω) and γ(ω) is continuous on I. Then J(ω) and −J(ω) are the one-way transfer functions which produce G(ω) on I, and there are no others.

Proof: The proof is similar to the proof of Theorem 1. We know that one of the solutions for the one-way transfer function is the function H(ω)=√{square root over (B(ω))}e^(jα(ω)), where β(ω)=2α(ω). Since G(ω)≠0 on I, H(ω) and J(ω) are nonzero on I. Then,

$\begin{matrix} {\frac{H(\omega)}{J(\omega)} = {\frac{\sqrt{B(\omega)}{\mathbb{e}}^{{j\alpha}{(\omega)}}}{\sqrt{B(\omega)}{\mathbb{e}}^{{j\gamma}{(\omega)}}} = {\mathbb{e}}^{j{\lbrack{{\alpha{(\omega)}} - {\gamma{(\omega)}}}\rbrack}}}} & (35) \end{matrix}$

By hypothesis γ(ω) is continuous on I and by Theorem 2 α(ω) is also continuous on I. Thus, α(ω)−γ(ω) is continuous on I. The conditions β(ω)=2α(ω)=2γ(ω) imply that for each ωεI, α(ω)−γ(ω) is either 0 or π. However, α(ω)−γ(ω) cannot switch between these two values without becoming discontinuous on I. Thus, either α(ω)−γ(ω)=0 for all ωεI, or α(ω)−γ(ω)=π for all ωεI. In the first case, we get J(ω)=H(ω), and in the second J(ω)=−H(ω).

Theorem 3 tells us that to get a one-way solution on any open interval I containing no zeros of the transfer function G(ω)=B (ω)e^(jβ(ω)), we simply form the function J(ω)=√{square root over (B(ω))}e^(jγ(ω)), choosing the values of γ(ω) satisfying β(ω)=2γ(ω) in such a way as to make the phase function γ(ω) continuous. Since it is known that there is a solution having this property, namely H(ω), it is always possible to do this.

Although the above theorems show how to reconstruct the two one-way transfer functions which generate the two-way function G(ω), they are useful only on a frequency interval I containing no zeros of G(ω). In general, G(ω) will be observed on a frequency interval (ω₁,ω₂) which may contain zeros. The following is a method that might get around this problem, assuming that there are only a finite number of zeros of G(ω) in (ω₁,ω₂), and that a one-way transfer function has derivatives of all orders on (ω₁,ω₂), not all of which are zero at any given frequency ω:

Let H(ω) be a one-way function that generates G(ω) on the interval (ω₁,ω₂), and assume that G (ω) has at least one zero on (ω₁,ω₂). The zeros of G(ω) will separate (ω₁,ω₂) into a finite number of abutting open frequency intervals J₁, J₂, . . . , J_(n). On each such interval the solution or H(ω) or −H(ω) will be found using either Theorem 1 or Theorem 3. We need to “stitch together” these solutions so that the stitched solution is either H(ω) or −H(ω) across all of (ω₁,ω₂). In order to do this, we need to know how to pair the solutions in two adjacent subintervals so that we aren't switching from H(ω) to −H(ω) or from −H(ω) to H(ω) in moving from one subinterval to the next.

We illustrate the stitching procedure starting with the first two adjacent open subintervals J₁ and J₂. These subintervals will abut at a frequency ω₁ which is a zero of G(ω) (of course, ω₁ is not contained in either subinterval). By our above assumption about the properties of a one-way transfer function, there must be a minimum positive integer n such that H^((n))(ω₁)≠0, where the superscript (n) denotes the n^(th) derivative. Then the limit of the n^(th) derivative of our one-way solution in J₁ as ω→ω₁ from the left will be either H^((n))(ω₁) or −H^((n))(ω₁) according to whether our solution in J₁ is H(ω) or −H(ω). Similarly, the limit of the n^(th) derivative of our one-way solution in J₂ as ω→ω₁ from the right will be either H^((n))(ω₁) or −H^((n))(ω₁) according to whether our solution in J₂ is H(ω) or −H(ω). Since H^((n))(ω₁)≠0, the two limits will be equal if and only if the solutions in J₁ and J₂ are both H(ω) or both −H(ω). If the left and right hand limits are unequal, we invert the solution in subinterval J₂. Otherwise, we don't.

After inverting the solution in subinterval J₂ (if necessary), we perform an identical procedure for subintervals J₂ and J₃, inverting the solution in subinterval J₃ (if necessary). Continuing in this fashion, we eventually build up a complete solution on the interval (ω₁,ω₂).

It would be desirable that high-order derivatives of H(ω) not be required in the above reconstruction procedure, since they are difficult to compute accurately in the presence of noise. This problem is unlikely to occur, because at any zero of G(ω) it seems very likely that the first derivative of H(ω) will be nonzero, and if not, very likely that the second derivative will be nonzero.

In a practical scheme, the two-way transfer function G(ω) will be measured at discrete frequencies, which must be close enough together to enable reasonably accurate computation of derivatives near the zeros of G(ω).

For RF-based distance measurements it is necessary to resolve an unknown number of closely spaced, overlapping, and noisy echoes of a ranging signal with a priori known shape. Assuming that ranging signal is a narrow-band, in frequency domain this RF phenomena can be described (modeled) as a sum of a number of sine waves, each per multipath component, and each with the complex attenuation and propagation delay of the path.

Taking the Fourier transform of the above mentioned sum will express this multipath model in the time domain. Exchanging the role of time and frequency variables in this time domain expression, this multipath model will become harmonic signals spectrum in which the propagation delay of the path is transformed to a harmonic signal.

The super (high) resolution spectral estimation methods are designed to distinguish closely-placed frequencies in the spectrum and used for estimating the individual frequencies of multiple harmonic signals, e.g., paths delays. As a result, path delays can be accurately estimated.

The super resolution spectral estimation makes use of the eigen-structure of the covariance matrix of the baseband ranging signal samples and covariance matrix intrinsic properties to provide a solution to an underlying estimation of individual frequencies, e.g. paths delays. One of the eigen-structure properties is that the eigenvalues can be combined and consequently divided into orthogonal noise and signal eigenvectors, aka subspaces. Another eigen-structure property is the rotation-invariant signal subspaces property.

The subspace decomposition technology (MUSIC, rootMUSIC, ESPRIT, etc.) relies on breaking the estimated covariance matrix of the observed data into two orthogonal subspaces, the noise subspace and the signal subspace. The theory behind the subspace decomposition methodology is that the projection of the observable onto the noise subspace consists of only the noise and the projection of the observable onto the signal subspace consists of only the signal.

The spectral estimation methods assume that signals are narrow-band, and the number of harmonic signals is also known, i.e., the size of the signal subspace needs to be known. The size of the signal subspace is called as the model size. In general, it cannot be known in any detail and can change rapidly—particularly indoors—as the environment changes. One of the most difficult and subtle issues when applying any subspace decomposition algorithm is the dimension of the signal subspace that can be taken as the number of frequency components present, and which is the number multipath reflections plus the direct path. Because of real-world measurement imperfections there always will be an error in the model size estimation, which in turn will result in loss of accuracy of frequencies estimation, i.e., distances.

To improve the distance measurement accuracy, one embodiment of the present invention includes six features that advance the state of the art in the methodology of subspace decomposition high resolution estimation. Included is combining two or more algorithms estimating individual frequencies by using different eigen-structure properties that further reduces the delay path determination ambiguity.

Root Music finds the individual frequencies, that when the observable is projected onto the noise subspace, minimizes the energy of the projection. The Esprit algorithm determines the individual frequencies from the rotation operator. And in many respects this operation is the conjugate of Music in that it finds the frequencies that, when the observable is projected onto the signal subspace, maximizes the energy of the projection.

The model size is the key to both of these algorithms, and in practice, in a complex signal environment such as seen in indoor ranging—the model size which provides the best performance for Music and Esprit are in general not equal, for reasons that will be discussed below.

For Music it is preferable to err on the side of identifying a basis element of the decomposition as a “signal eigen value” (Type I Error). This will minimize the amount of signal energy that is projected on the noise subspace and improve the accuracy. For Esprit—the opposite is true—it is preferable to err on the side of identifying a basis element of the decomposition as a “noise eigenvalue.” This is again a Type I Error. This will minimize the impact of noise on the energy projected onto the signal subspace. Therefore, the model size for Music will, in general, be somewhat larger than that for Esprit.

Secondly, in a complex signal environment, there arise occasions where, with the strong reflections and the potential that the direct path is in fact much weaker than some of the multipath reflections, the model size is difficult to estimate with sufficient statistical reliability. This issue is addressed by estimating a “base” model size for both Music and Esprit and the processing the observable data using Music and Esprit in a window of model sizes defined by the base model size for each. This results in multiple measurements for each measurement.

The first feature of the embodiment is the use of the F-statistic to estimate the model size (see above). The second feature is the use of different Type I Error probabilities in the F-statistic for Music and Esprit. This implements the Type I Error differences between Music and Esprit as discussed above. The third feature is the use of a base model size and a window in order to maximize the probability of detecting the direct path.

Because of the potentially rapidly changing physical and electronic environment, not every measurement will provide robust answers. This is addressed by using cluster analysis on multiple measurements to provide a robust range estimate. The fourth feature of the embodiment is the use of multiple measurements.

Because there are multiple signals present, the probability distribution of the multiple answers resulting from multiple measurements, each using multiple model sizes from both a Music and Esprit implementation, will be multimodal. Conventional cluster analysis will not be sufficient for this application. The fifth feature is the development of multimodal cluster analysis to estimate the direct range and equivalent range of the reflected multipath components. The sixth feature is the analysis of the statistics of the range estimates provided by the cluster analysis (range and standard deviation and combing those estimates that are statistically identical. This results in a more accurate range estimate.

The abovementioned method can be also used in wide bandwidth ranging signal location-finding systems.

Having thus described the different embodiments of a system and methods, it should be apparent to those skilled in the art that certain advantages of the described method and apparatus have been achieved. In particular, it should be appreciated by those skilled in the art that a system for tracking and locating objects can be assembled using FGPA or ASIC and standard signal processing software/hardware combination at a very small incremental cost. Such a system is useful in a variety of applications, e.g. locating people in indoor or in outdoor environments, harsh and hostile environments etc.

It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention. The invention is further defined by the following claims.

APPENDIX Derivation of r(t) in the Thresholded Method

Starting with expression (20), we obtain

$\begin{matrix} \begin{matrix} {{g(t)} = {\left( {a_{0} + {\sum\limits_{k = 1}^{M}\;{a_{k}\sin\mspace{11mu} k\;{{\pi\Delta}{ft}}}}} \right){{\sin\pi}\left( {{2N} + 1} \right)}{\Delta{ft}}}} \\ {= {{a_{0}\sin\;{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}} + {\sum\limits_{k = 1}^{M}\;{a_{k}\sin\;{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}\mspace{11mu}\sin\;{k{\pi\Delta}{ft}}}}}} \\ {= {{a_{0}\sin\;{\pi\left( {{2N} + 1} \right)}{\Delta{ft}}} + {\sum\limits_{k = 1}^{M}\;{\frac{1}{2}a_{k}\cos\;{\pi\left( {{2N} + 1 - k} \right)}{\Delta{ft}}}} -}} \\ {\sum\limits_{k = 1}^{M}\;{\frac{1}{2}a_{k}\cos\;{\pi\left( {{2N} + 1 + k} \right)}{\Delta{ft}}}} \\ {= {{a_{0}\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}} + {\sum\limits_{k = 1}^{M}\;{\frac{1}{2}a_{k}\cos\; 2{\pi\left( {N + \frac{1}{2} - \frac{k}{2}} \right)}{\Delta{ft}}}} -}} \\ {\sum\limits_{k = 1}^{M}\;{\frac{1}{2}a_{k}\cos\; 2{\pi\left( {N + \frac{1}{2} + \frac{k}{2}} \right)}{\Delta{ft}}}} \end{matrix} & ({A1}) \end{matrix}$

where the trigonometric identity sin x sin

$y = {{\frac{1}{2}{\cos\left( {x - y} \right)}} - {\frac{1}{2}{\cos\left( {x + y} \right)}}}$ is used.

Except for a₀, the coefficients a_(k) are zero for even k. The reason for this is that on the interval I, the function 1/sin πΔft that we are trying to approximate by h(t) is even about the center of I, but the basis functions sin kπΔft for even k, k≠0, are odd about the center of I, hence are orthogonal to 1/sin/πΔft on I. Thus, we can make the substitution k=2n+1 and let M be an odd positive integer. In fact, we will let M=2N+1. This choice has been experimentally determined to provide a good amount of cancellation of the oscillations in the interval I.

$\begin{matrix} {{g(t)} = {{a_{0}\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}} + {\sum\limits_{n = 0}^{N}\;{\frac{1}{2}a_{{2n} + 1}\cos\; 2{\pi\left( {N - n} \right)}{\Delta{ft}}}} - {\sum\limits_{n = 0}^{N}\;{\frac{1}{2}a_{{2n} + 1}\cos\; 2{\pi\left( {N + n + 1} \right)}{\Delta{ft}}}}}} & ({A2}) \end{matrix}$

Now we make the substitution k=N−n in the first summation and k=N+n+1 in the second summation to obtain

$\begin{matrix} \begin{matrix} \begin{matrix} {{g(t)} = {{a_{0}\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}} + {\sum\limits_{k = 0}^{N}\;{\frac{1}{2}a_{{2{({N - k})}} + 1}\cos\; 2\pi\; k\;{\Delta{ft}}}} -}} \\ {\sum\limits_{k = {N + 1}}^{{2N} + 1}\;{\frac{1}{2}a_{{2{({k - N})}} - 1}\cos\; 2\pi\; k\;{\Delta{ft}}}} \\ {= {{a_{0}\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}} + {\frac{1}{2}a_{{2N} + 1}} +}} \\ {{\sum\limits_{k = 1}^{N}\;{\frac{1}{2}a_{{2{({N - k})}} + 1}\cos\; 2\pi\; k\;{\Delta{ft}}}} - {\sum\limits_{k = {N + 1}}^{{2N} + 1}\;{\frac{1}{2}a_{{2{({k - N})}} - 1}\cos\; 2\pi\; k\;{\Delta{ft}}}}} \end{matrix} & ({A3}) \end{matrix} & \; \end{matrix}$

Subtracting g (t) from s (t) results in

$\begin{matrix} \begin{matrix} {{r(t)} = {{s(t)} - {g(t)}}} \\ {= {1 + {2{\sum\limits_{k = 1}^{N}\;{\cos\; 2\pi\; k\;{\Delta{ft}}}}} - {\frac{1}{2}a_{{2N} + 1}} -}} \\ {{\sum\limits_{k = 1}^{N}\;{\frac{1}{2}a_{{2{({N - k})}} + 1}\cos\; 2\pi\; k\;{\Delta{ft}}}} +} \\ {{\sum\limits_{k = {N + 1}}^{{2N} + 1}\;{\frac{1}{2}a_{{2{({k - N})}} - 1}\cos\; 2\pi\; k\;{\Delta{ft}}}} - {a_{0}\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}}} \end{matrix} & ({A4}) \end{matrix}$ Now let

$\begin{matrix} \begin{matrix} {b_{0} = {1 - {\frac{1}{2}a_{{2N} + 1}}}} \\ {{b_{k} = {{2 - {\frac{1}{2}a_{{2{({N - k})}} + 1}\mspace{14mu}{for}\mspace{14mu} k}} = 1}},2,\ldots\mspace{14mu},N} \\ {{b_{k} = {{\frac{1}{2}a_{{2{({k - N})}} - 1}\mspace{14mu}{for}\mspace{14mu} k} = {N + 1}}},{N + 2},\ldots\mspace{14mu},{{2N} + 1}} \\ {c = {- a_{0}}} \end{matrix} & ({A5}) \end{matrix}$

Then (A4) can be written as

$\begin{matrix} {{r(t)} = {b_{0} + {\sum\limits_{k = 1}^{{2N} + 1}\;{b_{k}\cos\; 2\pi\; k\;{\Delta{ft}}}} + {c\;\sin\; 2{\pi\left( {N + \frac{1}{2}} \right)}{\Delta{ft}}}}} & ({A6}) \end{matrix}$

The present invention relates to a positioning/locating method in wireless communication and other wireless networks that substantially obviate one or more of the disadvantages of the related art. The present invention advantageously improves the accuracy of tracking and locating functionality in multiple types of wireless network by utilizing multi-path mitigation processes, techniques and algorithms, described in co-pending application Ser. No. 12/502,809, These wireless networks include Wireless Personal Area Networks (WPGAN) such as ZigBee and Blue Tooth, wireless local area network (WLAN) such as WiFi and UWB, Wireless Metropolitan Area Networks, (WMAN) typically consisting of multiple WLANs, WiMax being the primary example, wireless Wide Area Networks (WAN) such as White Space TV Bands, and Mobile Devices Networks (MDN) that are typically used to transmit voice and data. MDNs are typically based Global System for Mobile Communications (GSM) and Personal Communications Service (PCS) standards. A more recent MDN is based on the Long Term Evolution (LTE) standard. These wireless networks are typically comprised of a combination of devices, including base stations, desktop, tablet and laptop computers, handsets, smartphones, actuators, dedicated tags, sensors as well as other communication and data devices (generally, all these devices are referred to as “wireless network devices”).

Existing location and positioning information solutions use multiple technologies and networks, including GPS, AGPS, Cell Phone Tower Triangulation, and Wi-Fi. Some of the methods used to derive this location information include RF Fingerprinting, RSSI, and TDOA. Although acceptable for E911 requirements, existing location and ranging methods do not have the reliability and accuracy required to support LBS and/or RTLS applications, especially indoors and urban environments.

The methods described in co-pending application Ser. No. 12/502,809 significantly improve the ability to accurately locate and track targeted devices within a single wireless network or a combination of multiple wireless networks. The exemplary embodiment is a significant improvement to the existing implementation of tracking and location methods used by wireless networks that use Enhanced Cell-ID and OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA).

Cell ID location technique allows estimating the position of the user (UE—User

Equipment) with the accuracy of the particular sector coverage area. Thus, the attainable accuracy depends on the cell (base station) sectoring scheme and antenna beam-width. In order to improve accuracy the Enhanced Cell ID technique adds RTT (Round Trip Time) measurements from the eNB. Note: Here, the RTT constitutes the difference between transmission of a downlink DPCH—Dedicated Physical Channel, (DPDCH)/DPCCH: Dedicated Physical Data Channel/Dedicated Physical Control Channel) frame and the beginning of a corresponding uplink physical frame. In this instance the abovementioned frame(s) act as a ranging signal. Based on the information of how long this signal propagates from eNB to the UE, the distance from eNB can be calculated (see FIG. 10).

In the Observed Time Difference of Arrival (OTDOA) technique the time of arrival of the signal coming from neighboring base stations (eNB) is calculated. The UE position can be estimated in the handset (UE-based method) or in the network (NT-based, UE-assisted method) once the signals from three base stations are received. The measured signal is the CPICH(Common Pilot Channel). The propagation time of signals is correlated with a locally generated replica. The peak of correlation indicates the observed time of propagation of the measured signal. Time difference of arrival values between two base stations determines a hyperbola. At least three reference points are needed to define two hyperbolas. The location of the UE is in the intersection of these two hyperbolas (see FIG. 11).

Idle Period Downlink (IPDL) is further OTDOA enhancement. The OTDOA-IPDL technique is based on the same measurements as the regular OTDOA Time measurements are taken during idle periods, in which serving eNB ceases its transmissions and allows the UE within the coverage of this cell to hear pilots coming from distant eNB(s). Serving eNB provides idle periods in continuous or burst mode. In the continuous mode, one idle period is inserted in every downlink physical frame (10 ms). In the burst mode, idle periods occur in a pseudo-random way. Further improvement is obtained via Time Aligned IPDL (TA-IPDL). Time alignment creates a common idle period, during which, each base station will either cease its transmission or transmit the common pilot. The pilot signal measurements will occur in idle period. There are several other techniques that may further enhance the DL OTDOA-IPDL method, for example Cumulative Virtual Blanking, UTDOA (Uplink TDOA), etc. All these techniques improve the ability to hear other (non-serving) eNB(s).

One significant drawback of the OTDOA based techniques is that the base stations timing relationships must be known, or measured (synchronized), for this method to be viable. For unsynchronized UMTS networks the 3GPP standard offers suggestion of how this timing may be recovered. However, networks operators are not implementing such solution. As a result, an alternative that uses the RTT measurements in lieu of the CPICH signal measurements was proposed (see U.S. Patent Publication No. 20080285505, John Carlson et al., SYSTEM AND METHOD FOR NETWORK TIMING RECOVERY IN COMMUNICATIONS NETWORKS).

All abovementioned methods/techniques are based on the terrestrial signals time of arrival and/or time difference of arrival measurements (RTT, CPICH, etc.). An issue with such measurements is that these are severely impacted by the multi-path. This, in turn, significantly degrades the abovementioned methods/techniques locate/track accuracy (see Jakub Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS).

One Multi-path mitigation technique uses detections/measurements from excess number of eNB(s) or Radio Base Stations (RBS). The minimum is three, but for multipath mitigation the number of RBS's required is at least six to eight (see METHOD AND ARRANGEMENT FOR DL-OTDOA (DOWNLINK OBSERVED TIME DIFFERENCE OF ARRIVAL) POSITIONING IN A LTE (LONG TERM EVOLUTION) WIRELESS COMMUNICATIONS SYSTEM, WO/2010/104436). However, the probability of an UE hearing from this large number of eNB(s) is much lower than from three eNB(s). This is because with large number of RBS (eNBs) there will be several ones that are far away from the UE and the received signal from these RBS(es) may fall below the UE receiver sensitivity level or the received signal will have low SNR.

In case of RF reflections (e.g., multi-path), multiple copies of the RF signal with various delay times are superimposed onto the DLOS (Direct Line Of Site) signal. Because CPICH, uplink DPCCH/DPDCH and other signals that are used in various CELL ID and OTDOA methods/techniques, including the RTT measurements, are of a limited bandwidth the DLOS signal and reflected signals cannot be differentiated without proper multi-path processing/mitigation; and without this multi-path processing these reflected signals will induce an error in the estimated time difference of arrival (TDOA) and time of arrival (TOA) measurements, including RTT measurements.

For example, 3 G TS 25.515 v.3.0.0 (199-10) standards define the RTT as “ . . . the difference between transmission of a downlink DPCH frame (signal) and the reception of the beginning (first significant path) of the corresponding uplink DPCCH/DPDCH frame (signal) from UE”. The standard does not define what constitutes this” first significant path”. The standard goes on noting that “The definition of the first significant path needs further elaboration”. For example, in heavy multipath environment it is a common occurrence whereby the DLOS signal, which is the first significant path, is severely attenuated (10 dB-20 dB) relatively to one or more reflected signal(s). If the “first significant path” is determined by measuring the signal strength, it may be one of the reflected signal(s) and not the DLOS signal. This will result in erroneous TOA/DTOA/RTT measurement(s) and loss of locating accuracy.

In prior wireless networks generations the locating accuracy was also impacted by the low sampling rate of frames (signals) that are used by the locate methods—RTT, CPCIH and other signals. The current third and following wireless network generations have much higher sampling rate. As a result, in these networks the locating accuracy real impact is from the terrestrial RF propagation phenomena (multipath).

The exemplary embodiment can be used in all wireless networks that employ reference and/or pilot signals, including simplex, half-duplex and full duplex modes of operation. For example, the exemplary embodiment operates with wireless networks that employ OFDM modulation and/or its derivatives. Thus, the exemplary embodiment operates with LTE networks.

It is also applicable to other wireless networks, including WiMax, WiFi, and White Space. Other wireless networks that do not use reference and/or pilot signals may employ one or more of the following types of alternate modulation embodiments of present invention as described in co-pending application Ser. No. 12/502,809: 1) where a portion of frame is dedicated to the ranging signal/ranging signal elements as described in co-pending application Ser. No. 12/502,809; 2) where the ranging signal elements (co-pending application Ser. No. 12/502,809) are embedded into transmit/receive signals frame(s); and 3) where the ranging signal elements (described in co-pending application Ser. No. 12/502,809) are embedded with the data.

These alternate embodiments employ multi-path mitigation processor and multi-path mitigation techniques/algorithms described in co-pending application Ser. No. 12/502,809 and can be used in all modes of operation: simplex, half-duplex and full duplex.

It is also likely that multiple wireless networks will, at the same time, utilize the preferred and/or alternate embodiments. By way of example, a smart phone can have Blue Tooth, WiFi, GSM and LTE functionality with the capability of operating on multiple networks at the same time. Depending on application demands and/or network availability, different wireless networks can be utilized to provide positioning/locating information.

The proposed exemplary embodiment method and system leverages the wireless network reference/pilot signals. Furthermore, the reference/pilot signal measurements might be combined with RTT (Round Trip Time) measurements or system timing. According to an exemplary embodiment, RF-based tracking and locating is implemented on 3GPP LTE cellular networks, but could be also implemented on other wireless networks, for example WiMax, Wi-Fi, LTE, sensors networks, etc. that employ a variety of signaling techniques. Both the exemplary and mentioned above alternative embodiments employ multi-path mitigation method/techniques and algorithms that are described in co-pending application Ser. No. 12/502,809 The proposed system can use software implemented digital signal processing.

The system of the exemplary embodiment leverages User Equipment (UE), e.g. cell phone or smart phone, hardware/software as well as Base Station (Node B)/enhanced Base Station (eNB) hardware/software. A base station generally consists of transmitters and receivers in a cabin or cabinet connected to antennas by feeders. These base stations include, Micro Cell, Pico Cell, Macro Cell, Umbrella Cell, Cell Phone towers, Routers and Femtocells. As a result, there will be little or no incremental cost to the UE device and overall system. At the same time the locate accuracy will be significantly improved.

The improved accuracy comes from the multipath mitigation that is provided by the present invention and the co-pending application Ser. No. 12/502,809. The exemplary embodiment uses multi-path mitigation algorithms, network reference signals and network node (eNB). These might be supplemented with RTT (Round Time Trip) measurements. The multi-path mitigation algorithms are implemented in UE and/or base station (eNB).

The exemplary embodiment advantageously uses the multi-path mitigation processor/algorithms (see co-pending application Ser. No. 12/502,809) that allow separating the DLOS signal and reflected signals, even when DLOS signal is significantly attenuated (10 dB-20 dB lower) relatively to one or more reflected signals. Thus, the exemplary embodiment significantly lowers the error in the estimated ranging signal DLOS time-of-flight and consequently TOA, RTT and DTOA measurements. The proposed multi-path mitigation and DLOS differentiating (recognizing) method can be used on all RF bands and wireless systems/networks. And it can support various modulation/demodulation techniques, including Spread Spectrum techniques, such as DSS (Direct Spread Spectrum) and FH(Frequency Hopping).

Additionally, noise reduction methods can be applied in order to further improve the method's accuracy. These noise reduction methods can include, but are not limited to, coherent summing, non-coherent summing, Matched filtering, temporal diversity techniques, etc. The remnants of the multi-path interference error can be further reduced by applying the post-processing techniques, such as, maximum likelihood estimation (e.g., Viterbi Algorithm), minimal variance estimation (Kalman Filter), etc.

In present invention exemplary embodiment the multi-path mitigation processor and multi-path mitigation techniques/algorithms do not change the RTT, CPCIH and other signals and/or frames. The present invention leverages wireless network reference signals that are used to obtain the channel response/estimation. The invention uses the channel estimation statistics that is generated by UE and/or eNB (see Iwamatsu et al., APPARATUS FOR ESTIMATING PROPAGATION PATH CHARACTERISTICS, US 2003/008156; U.S. Pat. No. 7,167,456 B2).

LTE networks use specific (non-data) reference signals (known signals) that are transmitted in every downlink subframe, and span entire cell bandwidth. The LTE reference signals are interspersed among LTE resource elements (see FIG. 9). From FIG. 2, reference signals (symbols) are transmitted every sixth subcarrier. Further, reference signals (symbols) are staggered in both time and frequency. In total, reference signals are covering every third subcarrier.

These reference signals are used for initial cell search by the UE, downlink signal strength measurements, scheduling and handover, etc. Included in the reference signals are UE-specific reference signals for channel estimation (response determination) for coherent demodulation. In addition to the UE-specific reference signals, other reference signals may be also used for channel estimation purposes, (see Chen et al., US patent publication No. 2010/0091826 A1).

LTE employs the OFDM (Orthogonal Frequency Division Multiplexing) modulation (technique). In LTE the ISI (Inter Symbol Interference) caused by multipath is handled by inserting Cyclic prefix (CP) at the beginning of each OFDM symbol. The CP provides enough delay so that delayed reflected signals of the previous OFDM symbol will die out before reaching the next OFDM symbol.

An OFDM symbol consists of multiple very tightly spaced subcarriers. Inside the OFDM symbol time-staggered copies of the current symbol (caused by multipath) result in Inter Carrier Interference (ICI). In LTE the ICI is handled (mitigated) by determining the multipath channel response and correcting the channel response in the receiver.

In LTE the multipath channel response (estimation) is computed in the receiver from subcarriers bearing the reference symbols. Interpolation is used to estimate the channel response on the remaining subcarriers. The channel response is calculated (estimated) in form of channel amplitude and phase. Once the channel response is determined (by periodic transmission of known reference signals), the channel distortion caused by multipath is mitigated by applying an amplitude and phase shift on a subcarrier-by-subcarrier basis (see Jim Zyren, Overview of the 3GPP Long Term Evolution Physical Layer, white paper).

LTE multipath mitigation is designed to remove the ISI and ICI, but not to separate the DLOS signal from reflected signals. For example, time-staggered copies of the current symbol make each modulated subcarrier signals spread in time, thus causing ICI. Correcting multipath channel response using the abovementioned LTE technique will shrink modulated subcarrier signals in time, but this type of correction does not guarantee that the resulting modulated subcarrier signals (inside the OFDM symbol) are DLOS signals. If DLOS modulated subcarrier signals are significantly attenuated relatively to delayed reflected signal(s), the resulting output signal will be the delayed reflected signal(s) and the DLOS signal will be lost.

In LTE compliant receiver, further signal processing includes DFT (Digital Fourier Transformation). It is well known that DFT technique(s) can resolve (remove) only copies of signal(s) that are delayed for times that are longer than or equal to the time that is inversely proportional to the signal and/or channel bandwidth. This method accuracy may be adequate for an efficient data transfer, but not accurate enough for precise distance measurement in a heavy multipath environment. For example, to achieve thirty meters accuracy, the signal and receiver channel bandwidths should be larger than or equal to ten megahertz ( 1/10 MHz=100 ns.). For better accuracy the signal and receiver channel bandwidths should be wider—one hundred megahertz for three meters.

However, CPICH, uplink DPCCH/DPDCH and other signals that are used in various CELL ID and OTDOA methods/techniques, including the RTT measurements, as well as the LTE received signal subcarriers have bandwidths that are significantly lower than ten megahertz. As a result, the currently employed (in LTE) method/technique will produce locating errors in the range of 100 meters.

To overcome the abovementioned limitations the invention uses a unique combination of implementations of subspace decomposition high resolution spectral estimation methodologies and multimodal cluster analysis. This analysis and related multi-path mitigation method/techniques and algorithms, described in the co-pending application Ser. No. 12/502,809, allow a reliable and accurate separation of DLOS path from other reflected signals paths.

Compared to methods/techniques used in the LTE, in a heavy multipath environment this method/techniques and algorithms (co-pending application Ser. No. 12/502,809) deliver 20× to 50× accuracy improvement in the distance measurement via reliable and accurate separation of DLOS path from other multi-path (MP) paths.

Methods/techniques and algorithms described in the co-pending application Ser. No. 12/502,809 require ranging signal complex amplitude estimation. Accordingly, the LTE reference signals used for channel estimation (response determination) can be also construed as a ranging signal in methods/techniques and algorithms described the co-pending application Ser. No. 12/502,809. In this case the ranging signal complex amplitude is the channel response that is calculated (estimated) by the LTE receiver in form of amplitude and phase. In other words, the channel response statistics that is calculated (estimated) by the LTE receiver can provide complex amplitude information that is required by the method/techniques and algorithms described in the co-pending application Ser. No. 12/502,809.

In ideal open space RF propagation environment with no multipath the phase change of the received signal (ranging signal), e.g. channel response phase, will be directly proportional to the signal's frequency (a straight line); and the RF signal time-of-flight (propagation delay) in such environment can be directly computed from the phase vs. frequency dependency by computing first derivative of the phase vs. frequency dependency. The result will be the propagation delay constant.

In this ideal environment the absolute phase value at initial (or any) frequency is not important because the derivative is not affected by the phase absolute values.

In a heavy multipath environment the received signal phase change vs. frequency is a complicated curve (not a straight line); and the first derivative does not provide information that could be used for accurate separation of DLOS path from other reflected signals paths. This is the reason for employing multipath mitigation processor and method(s)/techniques and algorithms described in the co-pending application Ser. No. 12/502,809.

If the phase and frequency synchronization (phase coherency) achieved in a given wireless network/system is very good, then multipath mitigation processor and method(s)/techniques and algorithms described in the co-pending application Ser. No. 12/502,809 will accurately separate DLOS path from other reflected signals paths and determine this DLOS path length (time-of-flight).

In this phase coherent network/system no additional measurements are required. In other words, one way ranging (simplex ranging) can be realized.

However, if the degree of synchronization (phase coherency) achieved in a given wireless network/system is not accurate enough, then in a heavy multipath environment the received signal phase and amplitude change vs. frequency might be very similar for measurements conducted at two or more different locations (distances). This phenomenon might lead to an ambiguity in received signal DLOS distance (time-of-flight) determination.

To resolve this ambiguity it is necessary to know the actual (absolute) phase value for at least one frequency.

However, the amplitude and phase vs. frequency dependency that is computed by the LTE receiver does not include an actual phase value because all amplitude and phase values are computed from the downlink reference signals, e.g. relative to each other. Thus, the amplitude and phase of the channel response that is calculated (estimated) by the LTE receiver needs actual phase value at least at one frequency (subcarrier frequency).

In LTE this actual phase value can be determined from one or more RTT measurement(s), TOA measurements; or from time-stamping of one or more received reference signals, provided that 1) these time stamps of transmitting these signals by eNB are also known at the receiver (or vice versa) and 2) the receiver and eNB clocks are well synchronized in time.

All of the above methods provide the time-of-flight values of one or more reference signals. From the time-of-flight values and frequencies of these reference signals actual phase values at one or more frequencies can be calculated.

The present invention achieves a highly accurate DLOS distance determination/locating in a heavy multipath environment by combining multi-path mitigation processor, method(s)/techniques and algorithms described in the co-pending application Ser. No. 12/502,809 with: 1) the amplitude and phase vs. frequency dependency that is computed by the LTE receiver or 2) a combination of the amplitude and phase vs. frequency dependency that is computed by the LTE receiver and actual phase value(s) for one or more frequencies obtained via RTT and/or TOA; and/or time-stamping measurements.

In these cases the actual phase value(s) is affected by the multipath. However, this does not impact the performance of methods/techniques and algorithms described in the co-pending application Ser. No. 12/502,809.

In LTE RTT/TOA/TDOA measurements can be carried out with the resolution of 5 meters. RTT measurements are carried during dedicated connections. Thus, multiple simultaneous measurements are possible when UE is in soft handover state, in which the DPCH frames are exchanged between the UE and different networks (base stations). Similar to RTT, TOA measurements provide the signal's time-of-flight (propagation delay), but TOA measurements cannot be made simultaneously (Jakub Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS).

In order to locate UE on plane DLOS distances have to be determined at least from/to three eNB(s). To locate UE in three-dimensional space minimum four DLOS distances from/to four eNB(s) would have to be determined (assuming that at least one eNB is not on the same plane).

An example of UE positioning method is shown in FIG. 1.

In case of very good synchronization RTT measurements are not required.

If the degree of synchronization is not accurate enough, then methods like OTDOA, Cell ID+RTT and others, for example AOA (Angle-of-Arrival) and its combinations with other methods, can be used for the UE locating.

The Cell ID+RTT track-locate method accuracy is impacted by the multipath (RTT measurements) and the eNB (base station) antenna beamwidth. Base stations antennas beamwidths are between 33 and 65 degrees. These wide beamwidths results in locating error of 50-150 meters in urban areas (Jakub Marek Borkowski: Performance of Cell ID+RTT Hybrid Positioning Method for UMTS). Considering that in a heavy multipath environment the current LTE RTT distance measurement average error is approximately 100 meters, the overall expected average locate error of the currently employed by LTE Cell ID+RTT method is approximately 150 meters.

One of the embodiments of the present invention is the UE locating based on the AOA method, whereby one or more reference signals from the UE is used for the UE locate purposes. It involves an AOA determination device location for determining the DLOS AOA. The device can be collocated with the base station and/or installed at another one or more locations independent from the base station location. The coordinates of these locations are presumably known. No changes are required on the UE side.

This device includes a small antenna array and is based on a variation of the same multipath mitigation processor, method(s)/techniques and algorithms described in the co-pending application Ser. No. 12/502,809. This one possible embodiment of the present invention advantage is precise determination (very narrow beamwidth) of the AOA of the DLOS RF energy from an UE unit.

In one other option this added device is receive only device. As a result, its size/weight and cost are very low.

The combination of the present invention exemplary embodiment in which accurate DLOS distance measurements are obtained and one of the above-mentioned embodiments of the present invention in which an accurate DLOS AOA determination can be made will greatly improve the Cell ID+RTT track-locate method precision—10× or greater. Another advantage of this approach is that the UE location can be determined at any moment, (does not require placing UE in soft handover mode).

Another option of determining the DLOS AOA is to use the existing eNB antenna array and the eNB equipment. This option may further lower the cost of implementation of the improved Cell ID+RTT method. However, because eNB antennas are not designed for the locating applications, the positioning accuracy may be degraded. Also, network operators may be unwilling to implement required changes in base station (software/hardware). 

1. A system for tracking and locating wireless network devices, the system comprising: a multipath mitigation processor configured to receive and process reference signals of the wireless network; the multipath mitigation processor using the reference signals to measure a distance between the wireless network devices; wherein a high-resolution spectrum estimation analysis is used to reduce spatial ambiguity in the distance measurement, including estimating a model size for a number of frequency components, and calculating the distance based on a distribution of the frequency components; wherein the system employs noise reduction methods that include any of coherent summing, non-coherent summing, matched filtering and temporal diversity; and wherein a remainder of the multi-path interference error is further reduced by applying the post-processing techniques, including any of maximum likelihood estimation (Viterbi Algorithm) and minimal variance estimation (Kalman Filter).
 2. The system of claim 1, wherein the multipath mitigation processor is configured to receive RTT (Round Trip Time) measurement(s) from a wireless network device to compute an actual (absolute) phase value or phase and amplitude values of the reference signals of the wireless network.
 3. The system of claim 1, wherein the multipath mitigation processor is configured to receive TOA (Time of Arrival) measurements and to compute phase and amplitude values of the reference signals of the wireless network.
 4. The system of claim 1, wherein the multipath mitigation processor is configured to receive DTOA (Differential Time of Arrival) and to compute an actual (absolute) phase value of the reference signals of the wireless network.
 5. The system of claim 1, wherein the multipath mitigation processor time-stamps and processes the reference signals so as to compute an actual (absolute) phase value or phase and amplitude values of the reference signals of the wireless network, for determining the distance.
 6. The system of claim 1, wherein the multipath mitigation processor uses a channel estimation (response determination) computed by the wireless network devices, for determining the distance.
 7. The system of claim 1, wherein the wireless receiver is configured to receive narrow bandwidth signals.
 8. The system of claim 1, wherein the multipath mitigation processor supports half-duplex, full duplex or simplex mode of operation.
 9. The system of claim 1, wherein the multipath mitigation processor uses any of the following methods to estimate the distance and track-locate: Time of Arrival (TOA); Differential Time of Arrival (DTOA); a combination of TOA and DTOA; Triangulation and/or Trilateration; Virtual triangulation and Reverse Virtual Triangulation; OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA); Enhanced Cell ID; CELL ID+RTT; Cell ID+AOA (Angle of Arrival); and Cell ID+AOA (Angle of Arrival)+RTT.
 10. A method for tracking and locating wireless network devices, the method comprising: configuring a multipath mitigation processor to receive and process a ranging signal of the wireless network; configuring a multipath mitigation processor to generate and transmit a ranging signal of the wireless network; using the ranging signal and the processor to measure a distance between the wireless network devices; and wherein a high-resolution spectrum estimation analysis is used to reduce spatial ambiguity in the distance measurement, including estimating a model size for a number of frequency components, and calculating the distance based on a distribution of the frequency components; and wherein the system employs noise reduction methods that include any of coherent summing, non-coherent summing, matched filtering and temporal diversity; and wherein a remainder of the multi-path interference error is further reduced by applying the post-processing techniques, including any of maximum likelihood estimation (Viterbi Algorithm) and minimal variance estimation (Kalman Filter).
 11. The method of claim 10, wherein a portion of a transmit/receive wireless network frame is dedicated to ranging signal components.
 12. The method of claim 10, wherein the ranging signal components are spread across (embedded) the transmit/receive frame.
 13. The method of claim 10, wherein the ranging signal components are embedded with the transmit/receive data.
 14. The method of claim 10, wherein the ranging signal is a wide bandwidth signal and the signal comprises of different individual narrow bandwidth frequency components.
 15. The method of claim 10, wherein the ranging signal is a narrow bandwidth signal, and the system uses different individual narrow bandwidth frequency components to generate the narrow bandwidth ranging signal and a base-band narrow bandwidth ranging signal.
 16. The method of claim 10, wherein different narrow bandwidth frequency components of the ranging signal and a base-band ranging signal are pseudorandomly selected, the components are contiguous and spaced apart in frequency.
 17. The method of claim 10, wherein the reference signal is a pilot signal.
 18. A system for tracking and locating wireless network devices, the system comprising: a multipath mitigation processor configured to receive and process a pilot signal of the wireless network; and means for using the pilot signal and the processor to measure a distance between the wireless network devices by utilizing a high-resolution spectrum estimation analysis to reduce spatial ambiguity in the distance measurement, including estimating a model size for a number of frequency components, and calculating the distance based on a distribution of the frequency components; and wherein the system employs noise reduction methods that include any of coherent summing, non-coherent summing, matched filtering and temporal diversity; and wherein a remainder of the multi-path interference error is further reduced by applying the post-processing techniques, including any of maximum likelihood estimation (Viterbi Algorithm) and minimal variance estimation (Kalman Filter).
 19. The system of claim 18, wherein the reference signal is a pilot signal.
 20. The system of claim 18, further comprising a wireless receiver configured to receive a narrow bandwidth signal as the pilot signal.
 21. The system of claim 18, wherein the multipath mitigation processor supports half-duplex, full duplex or simplex mode of operation.
 22. The system of claim 18, wherein the multipath mitigation processor uses any of the following methods to estimate the distance and track-locate: Time of Arrival (TOA); Differential Time of Arrival (DTOA); a combination of TOA and DTOA; Triangulation and/or Trilateration; Virtual triangulation and Reverse Virtual Triangulation; OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA); Enhanced Cell ID; CELL ID+RTT; Cell ID+AOA (Angle of Arrival); and Cell ID+AOA (Angle of Arrival)+RTT.
 23. A system for tracking and locating wireless network devices, comprising: a wireless receiver adapted to interface to wireless network devices to measure the Angle of Arrival (AOA) from a network device; wherein the wireless receiver is configured to receive signals from wireless network devices; wherein the wireless receiver is configured to interface to a multipath mitigation processor; wherein the received signals are used for measuring the AOA from a wireless network c device, and the multipath mitigation processor configured and process the received signals; wherein a high-resolution spectrum estimation analysis is used to reduce spatial ambiguity in the distance measurement, including estimating a model size for a number of multi-path frequency components, and calculating the distance based on a distribution of the frequency components; and wherein the system employs noise reduction methods that include any of coherent summing, non-coherent summing, matched filtering and temporal diversity; and wherein the system further reduces a remainder of the multi-path interference error by applying the post-processing techniques, including any of maximum likelihood estimation (Viterbi Algorithm) and minimal variance estimation (like Kalman Filter).
 24. The system of claim 23, wherein the multipath mitigation processor uses any of the following methods to estimate the distance and track-locate: Time of Arrival (TOA); Differential Time of Arrival (DTOA); a combination of TOA and DTOA; Triangulation and/or Trilateration; Virtual triangulation and Reverse Virtual Triangulation; OTDOA (Observed Time Difference of Arrival), including DL-OTDOA (Downlink OTDOA); Enhanced Cell ID; CELL ID+RTT; Cell ID+AOA (Angle of Arrival); and Cell ID+AOA (Angle of Arrival)+RTT. 